ADL5902 Datasheet by Analog Devices Inc.

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ANALOG DEVICES
50 MHz to 9 GHz
65 dB TruPwr Detector
Data Sheet ADL5902
Rev. B Document Feedback
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responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
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Technical Support www.analog.com
FEATURES
Accurate rms-to-dc conversion from 50 MHz to 9 GHz
Single-ended input dynamic range of 65 dB
No balun or external input matching required
Waveform and modulation independent, such as
GSM/CDMA/W-CDMA/TD-SCDMA/WiMAX/LTE
Linear-in-decibels output, scaled 53 mV/dB
Transfer function ripple: <±0.1 dB
Temperature stability: <±0.3 dB
All functions temperature and supply stable
Operates from 4.5 V to 5.5 V from −40°C to +125°C
Power-down capability to 1.5 mW
Pin-compatible with the 50 dB dynamic range AD8363
APPLICATIONS
Power amplifier linearization/control loops
Transmitter power controls
Transmitter signal strength indication (TSSI)
RF instrumentation
FUNCTIONAL BLOCK DIAGRAM
08218-001
X
2
X
2
BIAS AND POWER-
DOWN CONTROL
1
NC
NC
NC
LINEAR-IN-dB VGA
(NEGATIVE SLOPE)
I
DET
26pF
2
3
4
11
10
9
5
6
7
8
16
15
14
13
ADL5902
12
VREF
2.3V
TEMPERATURE
SENSOR
INLO
INHI
V
POS POS
TEMP
VSET
VOUT
CLPF
COMMCOMMVTGTVREFTADJ/PWDN
G = 5
I
TGT
Figure 1.
GENERAL DESCRIPTION
The ADL5902 is a true rms responding power detector that has
a 65 dB measurement range when driven with a single-ended
50  source. This feature makes the ADL5902 frequency
versatile by eliminating the need for a balun or any other form
of external input tuning for operation up to 9 GHz.
The ADL5902 provides a solution in a variety of high frequency
systems requiring an accurate measurement of signal power.
Requiring only a single supply of 5 V and a few capacitors, it is
easy to use and capable of being driven single-ended or with a
balun for differential input drive. The ADL5902 can operate
from 50 MHz to 9 GHz and can accept inputs from −62 dBm to
at least +3 dBm with large crest factors, such as GSM, CDMA,
W-CDMA, TD-SCDMA, WiMAX, and LTE modulated signals.
The ADL5902 can determine the true power of a high
frequency signal having a complex low frequency modulation
envelope or can be used as a simple low frequency rms
voltmeter. Used as a power measurement device, VOUT is
connected to VSET. The output is then proportional to the
logarithm of the rms value of the input. In other words, the reading
is presented directly in decibels and is scaled 1.06 V per decade,
or 53 mV/dB; other slopes are easily arranged. In controller mode,
the voltage applied to VSET determines the power level required at
the input to null the deviation from the set point. The output
buffer can provide high load currents.
The ADL5902 has 1.5 mW power consumption when powered
down by a logic high applied to the PWDN pin. It powers up
within approximately 5 µs to the nominal operating current of
73 mA at 25°C. The ADL5902 is supplied in a 4 mm × 4 mm,
16-lead LFCSP for operation over the wide temperature range
of −40°C to +125°C.
The ADL5902 is also pin-compatible with the AD8363, 50 dB
dynamic range TruPwr™ detector. This feature allows the
designer to create one circuit layout for projects requiring
different dynamic ranges. A fully populated RoHS-compliant
evaluation board is available.
ADL5902 Data Sheet
Rev. B | Page 2 of 28
TABLE OF CONTENTS
Features .............................................................................................. 1
Applications ....................................................................................... 1
Functional Block Diagram .............................................................. 1
General Description ......................................................................... 1
Revision History ............................................................................... 2
Specifications ..................................................................................... 3
Absolute Maximum Ratings ............................................................ 7
ESD Caution .................................................................................. 7
Pin Configuration and Function Descriptions ............................. 8
Typical Performance Characteristics ............................................. 9
Theory of Operation ...................................................................... 15
Square Law Detector and Amplitude Target .............................. 15
RF Input Interface ...................................................................... 16
Small Signal Loop Response ..................................................... 17
Temperature Sensor Interface ................................................... 17
VREF Interface ........................................................................... 17
Temperature Compensation Interface ..................................... 17
Power-Down Interface ............................................................... 18
VSET Interface ............................................................................ 18
Output Interface ......................................................................... 18
VTGT Interface .......................................................................... 19
Basis for Error Calculations ...................................................... 19
Measurement Mode Basic Connections.................................. 19
Setting VTADJ .................................................................................. 20
Setting VTGT ................................................................................. 20
Choosing a Value for CLPF ............................................................ 20
Output Voltage Scaling .............................................................. 23
System Calibration and Error Calculation .............................. 24
High Frequency Performance ................................................... 25
Low Frequency Performance .................................................... 25
Description of Characterization ............................................... 25
Evaluation Board Schematics and Artwork ................................ 26
Assembly Drawings .................................................................... 27
Outline Dimensions ....................................................................... 28
Ordering Guide .......................................................................... 28
REVISION HISTORY
8/2016—Rev. A to Rev. B
Changes to Figure 2 .......................................................................... 8
Updated Outline Dimensions ....................................................... 28
Changes to Ordering Guide .......................................................... 28
7/2011—Rev. 0 to Rev. A
Updated Format .................................................................. Universal
Changes to Measurement Mode Basic Connections Section and
Figure 45 .......................................................................................... 19
Changes to Setting VTGT Section and Choosing a Value for
CLPF Section ...................................................................................... 20
Changes to Output Voltage Scaling Section, Figure 49, and
Table 7 .............................................................................................. 23
Changes to Figure 54 and Table 8 ................................................. 26
Changes to Figure 55 and Figure 56 ............................................. 27
4/2010—Revision 0: Initial Version
Data Sheet ADL5902
Rev. B | Page 3 of 28
SPECIFICATIONS
VS = 5 V, TA = 25°C, ZO = 50 Ω, single-ended input drive, RT = 60.4 Ω, VOUT connected to VSET, VTGT = 0.8 V, CLPF = 0.1 µF. Negative
current values imply that the ADL5902 is sourcing current out of the indicated pin.
Table 1.
Parameter Test Conditions/Comments Min Typ Max Unit
OVERALL FUNCTION
Frequency Range 50 to 9000 MHz
RF INPUT INTERFACE Pins INHI, INLO, ac-coupled
Input Impedance Single-ended drive, 50 MHz 2000 Ω
Common Mode Voltage 2.5 V
100 MHz
±1.0 dB Dynamic Range CW input, TA = +25°C, VTADJ = 0.5 V 63 dB
Maximum Input Level, ±1.0 dB Calibration at −60 dBm, −45 dBm, and 0 dBm 3 dBm
Minimum Input Level, ±1.0 dB Calibration at −60 dBm, −45 dBm, and 0 dBm −60 dBm
Deviation vs. Temperature Deviation from output at 25°C
−40°C < TA < +85°C; PIN = 0 dBm −0.11/+0.25 dB
−40°C < TA < +85°C; PIN = −45 dBm −0.22/+0.15 dB
−40°C < TA < +125°C; PIN = 0 dBm −0.35/+0.25 dB
−40°C < TA < +125°C; PIN = −45 dBm −0.22/+0.15 dB
Logarithmic Slope −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
53.8 mV/dB
Logarithmic Intercept −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
−62.1 dBm
700 MHz
±1.0 dB Dynamic Range CW input, TA = +25°C,VTADJ = 0.4 V 61 dB
Maximum Input Level, ±1.0 dB Calibration at −60 dBm, −45 dBm, and 0 dBm 1 dBm
Minimum Input Level, ±1.0 dB Calibration at −60 dBm, −45 dBm, and 0 dBm −60 dBm
Deviation vs. Temperature Deviation from output at 25°C
−40°C < TA < +85°C; PIN = 0 dBm +0.3/−0.2 dB
−40°C < TA < +85°C; PIN = −45 dBm −0.1/0 dB
−40°C < TA < +125°C; PIN = 0 dBm +0.3/−0.4 dB
−40°C < TA < +125°C; PIN = −45 dBm −0.1/0 dB
Logarithmic Slope −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
53.7 mV/dB
Logarithmic Intercept −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
−62.8 dBm
900 MHz
±1.0 dB Dynamic Range CW input, TA = +25°C, VTADJ = 0.4 V 61 dB
Maximum Input Level, ±1.0 dB Calibration at −60 dBm, −45 dBm, and 0 dBm 1 dBm
Minimum Input Level, ±1.0 dB Calibration at −60 dBm, −45 dBm, and 0 dBm −60 dBm
Deviation vs. Temperature Deviation from output at 25°C
−40°C < TA < +85°C; PIN = 0 dBm +0.3/−0.2 dB
−40°C < TA < +85°C; PIN = −45 dBm 0/−0.1 dB
−40°C < TA < +125°C; PIN = 0 dBm +0.3/−0.4 dB
−40°C < TA < +125°C; PIN = −45 dBm 0/−0.1 dB
Logarithmic Slope −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
53.7 mV/dB
Logarithmic Intercept −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
−62.7 dBm
ADL5902 Data Sheet
Rev. B | Page 4 of 28
Parameter Test Conditions/Comments Min Typ Max Unit
Deviation from CW Response 11.02 dB peak-to-rms ratio (CDMA2000) 0.1 dB
5.13 dB peak-to-rms ratio (16 QAM) 0.05 dB
2.76 dB peak-to-rms ratio (QPSK) 0.05 dB
1.9 GHz
±1.0 dB Dynamic Range CW input, TA = +25°C, VTADJ = 0.4 V 64 dB
Maximum Input Level, ±1.0 dB Calibration at −60 dBm, −45 dBm, and 0 dBm 3 dBm
Minimum Input Level, ±1.0 dB Calibration at −60 dBm, −45 dBm, and 0 dBm −61 dBm
Deviation vs. Temperature Deviation from output at 25°C
−40°C < TA < +85°C; PIN = 0 dBm −0.1/0 dB
−40°C < TA < +85°C; PIN = −45 dBm −0.3/+0.3 dB
−40°C < TA < +125°C; PIN = 0 dBm −0.1/0 dB
−40°C < TA < +125°C; PIN = −45 dBm −0.3/+0.4 dB
Logarithmic Slope −45 dBm < PIN < 0 dBm; calibration at −45 dBm,
and 0 dBm
52.6 mV/dB
Logarithmic Intercept −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
−62.6 dBm
2.14 GHz
±1.0 dB Dynamic Range CW input, TA = +25°C, VTADJ = 0.4 V 65 dB
Maximum Input Level, ±1.0 dB Calibration at −60 dBm, −45 dBm, and 0 dBm 3 dBm
Minimum Input Level, ±1.0 dB Calibration at −60 dBm, −45 dBm, and 0 dBm −62 dBm
Deviation vs. Temperature Deviation from output at 25°C
−40°C < TA < +85°C; PIN = 0 dBm −0.1/0 dB
−40°C < TA < +85°C; PIN = −45 dBm −0.3/+0.3 dB
−40°C < TA < +125°C; PIN = 0 dBm −0.1/0 dB
−40°C < TA < +125°C; PIN = −45 dBm −0.3/+0.4 dB
Logarithmic Slope −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
52.4 mV/dB
Logarithmic Intercept −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
−62.9 dBm
Deviation from CW Response 12.16 dB peak-to-rms ratio (four-carrier W-CDMA) −0.1 dB
11.58 dB peak-to-rms ratio (LTE TM1 1CR 20 MHz BW) −0.1 dB
10.56 dB peak-to-rms ratio (one-carrier W-CDMA) −0.1 dB
6.2 dB peak-to-rms ratio (64 QAM) −0.07 dB
2.6 GHz
±1.0 dB Dynamic Range CW input, TA = +25°C, VTADJ = 0.45 V 65 dB
Maximum Input Level, ±1.0 dB Calibration at −60, −45 and 0 dBm 5 dBm
Minimum Input Level, ±1.0 dB Calibration at −60, −45 and 0 dBm −60 dBm
Deviation vs. Temperature Deviation from output at 25°C
−40°C < TA < +85°C; PIN = 0 dBm 0.4/0 dB
−40°C < TA < +85°C; PIN = −45 dBm +0.5/−0.6 dB
−40°C < TA < +125°C; PIN = 0 dBm 0.6/0 dB
−40°C < TA < +125°C; PIN = −45 dBm +0.7/−0.6 dB
Logarithmic Slope −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
51.0 mV/dB
Logarithmic Intercept −45 dBm < PIN < 0 dBm; calibration at −45 dBm
and 0 dBm
−62.1 dBm
3.5 GHz
±1.0 dB Dynamic Range CW input, TA = +25°C, VTADJ = 0.5 V 57 dB
Maximum Input Level, ±1.0 dB Calibration at −60 dBm, −40 dBm, and 0 dBm 8 dBm
Minimum Input Level, ±1.0 dB Calibration at −60 dBm, −40 dBm, and 0 dBm −49 dBm
Data Sheet ADL5902
Rev. B | Page 5 of 28
Parameter Test Conditions/Comments Min Typ Max Unit
Deviation vs. Temperature Deviation from output at 25°C
−40°C < TA < +85°C; PIN = 0 dBm 0.2/0 dB
−40°C < TA < +85°C; PIN = −40 dBm −0.2/+0.4 dB
−40°C < TA < +125°C; PIN = 0 dBm +0.2/−0.3 dB
−40°C < TA < +125°C; PIN = −40 dBm −0.2/+0.4 dB
Logarithmic Slope −40 dBm < PIN < 0 dBm; calibration at −30 dBm
and 0 dBm
49.6 mV/dB
Logarithmic Intercept −40 dBm < PIN < 0 dBm; calibration at −30 dBm
and 0 dBm
−63.1 dBm
5.8 GHz
±1.0 dB Dynamic Range CW input, TA = +25°C, VTADJ = 0.95 V 61 dB
Maximum Input Level, ±1.0 dB Calibration at −50 dBm, −30 dBm, and 0 dBm 9 dBm
Minimum Input Level, ±1.0 dB Calibration at −50 dBm, −30 dBm, and 0 dBm −52 dBm
Deviation vs. Temperature Deviation from output at 25°C
−40°C < TA < +85°C; PIN = 0 dBm −0.8/0 dB
−40°C < TA < +85°C; PIN = −30 dBm −1.3/+0.1 dB
−40°C < TA < +125°C; PIN = 0 dBm −1.6/0 dB
−40°C < TA < +125°C; PIN = −30 dBm −1.3/+0.1 dB
Logarithmic Slope −30 dBm < PIN < 0 dBm; calibration at −30 dBm
and 0 dBm
42.7 mV/dB
Logarithmic Intercept −30 dBm < PIN < 0 dBm; calibration at −30 dBm
and 0 dBm
−54.1 dBm
OUTPUT INTERFACE VOUT (Pin 6)
Output Swing, Controller Mode Swing range minimum, RL ≥ 500 Ω to ground 0.03 V
Swing range maximum, RL ≥ 500 Ω to ground 4.8 V
Current Source/Sink Capability 10/10 mA
Voltage Regulation ILOAD = 8 mA, source/sink +0.2/−0.2 %
Output Noise RFIN = 2.14 GHz, −20 dBm, fNOISE = 100 kHz,
CLPF = 220 pF
25 nV/√Hz
Rise Time Transition from no input to 1 dB settling at
PIN = −10 dBm, CLPF = 220 pF
3 μs
Fall Time Transition from −10 dBm to off (1 dB of final value),
CLPF = 220 pF
25 μs
SETPOINT INPUT VSET (Pin 7)
Voltage Range Log conformance error ≤ 1 dB, minimum 2.14 GHz 3.5 V
Log conformance error ≤ 1 dB, maximum 2.14 GHz 0.23 V
Input Resistance 72
Logarithmic Scale Factor f = 2.14 GHz 52.4 mV/dB
Logarithmic Intercept f = 2.14 GHz −62.9 dBm
TEMPERATURE COMPENSATION Pin TADJ/PWDN (Pin 1)
Input Voltage Range 0 VS V
Input Bias Current VTADJ = 0.4 V 2 μA
Input Resistance VTADJ = 0.4 V 200
VOLTAGE REFERENCE VREF (Pin 11)
Output Voltage PIN = −55 dBm 2.3 V
Temperature Sensitivity 25°C ≤ TA ≤ 125°C −0.16 mV/°C
−15°C TA ≤ +25°C 0.045 mV/°C
−40°C TA ≤ −15°C −0.04 mV/°C
Short-Circuit Current Source/ Sink
Capability
25°C ≤ TA ≤ 125°C 4/0.05 mA
−40°C TA < +25°C 3/0.05 mA
Voltage Regulation TA = 25°C, ILOAD = 2 mA −0.4 %
ADL5902 Data Sheet
Rev. B | Page 6 of 28
Parameter Test Conditions/Comments Min Typ Max Unit
TEMPERATURE REFERENCE TEMP (Pin 8)
Output Voltage TA = 25°C, RL ≥ 10 kΩ 1.4 V
Temperature Coefficient −40°C ≤ TA ≤ +125°C, RL ≥ 10 kΩ 4.9 mV/°C
Short-Circuit Current Source/ Sink
Capability
25°C ≤ TA ≤ 125°C 4/0.05 mA
−40°C TA < +25°C 3/0.05 mA
Voltage Regulation TA = 25°C, ILOAD = 1 mA −2.8 %
RMS TARGET INTERFACE VTGT (Pin 12)
Input Voltage Range 0.2 2.5 V
Input Bias Current VTGT = 0.8 V 8 μA
Input Resistance 100
POWER-DOWN INTERFACE Pin TADJ/PWDN (Pin 1)
Voltage Level to Enable VPWDN decreasing 4 V
Voltage Level to Disable VPWDN increasing 4.9 V
Input Current VPWDN = 5 V 1 μA
V
PWDN = 4.5 V 500 μA
V
PWDN = 0 V 3 μA
Enable Time VTADJ low to VOUT at 1 dB of final value,
CLPA/B = 220 pF, PIN = 0 dBm
5 μs
Disable Time VTADJ high to VOUT at 1 dB of final value,
CLPA/B = 220 pF, PIN = 0 dBm
3 μs
POWER SUPPLY INTERFACE VPOS (Pin 3, Pin 10)
Supply Voltage 4.5 5 5.5 V
Quiescent Current TA = 25°C, PIN < −60 dBm 73 mA
T
A = 125°C, PIN < −60 dBm 90 mA
Power-Down Current VTADJ > VS − 0.1 V 300 μA
Am ESD (elemosmic dismarge) sen ve device Chavged dewces and (1mm boavds (an dlsthavge wmwm daemon Ahhough cm: pvodud feamves pnlcmcd or propncmvy pmmcuon (Nanny, damage may 0((uv on dame: subjeued xo hlgh enevgy ESD Thevefove. pvopev ESD pvecawons should be \aken w avmd performante degradation or ‘0); cl mnmnamy
Data Sheet ADL5902
Rev. B | Page 7 of 28
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Rating
Supply Voltage, VPOS 5.5 V
Input Average RF Power1 21 dBm
Equivalent Voltage, Sine Wave Input 2.51 V p-p
Internal Power Dissipation 550 mW
θJC2 10.6°C/W
θJB2 35.3°C/W
θJA2 57.2°C/W
ΨJT2 1.0°C/W
ΨJB2 34°C/W
Maximum Junction Temperature 150°C
Operating Temperature Range −40°C to +125°C
Storage Temperature Range −65°C to +150°C
Lead Temperature (Soldering, 60 sec) 300°C
1 This is for long durations. Excursions above this level, with durations much
less than 1 second, are possible without damage.
2 No airflow with the exposed pad soldered to a 4-layer JEDEC board.
Stresses at or above those listed under Absolute Maximum
Ratings may cause permanent damage to the product. This is a
stress rating only; functional operation of the product at these
or any other conditions above those indicated in the operational
section of this specification is not implied. Operation beyond
the maximum operating conditions for extended periods may
affect product reliability.
ESD CAUTION
ADL5902 Data Sheet
Rev. B | Page 8 of 28
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
08218-002
NOTES
1. NC = NO CONNECT.
2. THE EXPOSED PAD IS COMM AND SHOULD
HAVE BOTH A GOOD THERMAL AND GOOD
ELECTRICAL CONNECTION TO GROUND.
T
ADJ/PWDN
NC
VPOS
COMM
VREF
VTGT
VPOS
COMM
CLPF
VOUT
VSET
TEMP
INLO
NC
INHI
NC
PIN 1
INDICATOR
12
11
10
1
3
49
2
6
5
7
8
16
15
14
13
ADL5902
TOP VIEW
(Not to Scale)
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No. Mnemonic Description
1 TADJ/PWDN
This is a dual function pin used for controlling the amount of nonlinear intercept temperature compensation at
voltages <2.5 V and/or for shutting down the device at voltages >4 V. If the shutdown function is not used, this pin
can be connected to the VREF pin through a voltage divider. See Figure 41 for an equivalent circuit.
2 NC No Connect. Do not connect this pin.
3, 10 VPOS Supply for the Device. Connect this pin to a 5 V power supply. Pin 3 and Pin 10 are not internally connected;
therefore, both must connect to the source.
4, 9, EPAD COMM System Common Connection. Connect these pins via low impedance to system common. The exposed paddle
is also COMM and must have both a good thermal and good electrical connection to ground.
5 CLPF
Connection for RMS Averaging Capacitor. Connect a ground-referenced capacitor to this pin. A resistor can be
connected in series with this capacitor to modify loop stability and response time. See Figure 43 for an
equivalent circuit.
6 VOUT
Output. In measurement mode, this pin is connected to VSET. In controller mode, this pin can drive a gain
control element. See Figure 43 for an equivalent circuit.
7 VSET
The voltage applied to this pin sets the decibel value of the required RF input voltage that results in zero
current flow in the loop integrating capacitor pin, CLPF. This pin controls the variable gain amplifier (VGA) gain
such that a 50 mV change in VSET changes the gain by approximately 1 dB. See Figure 42 for an equivalent circuit.
8 TEMP Temperature Sensor Output of 1.4 V at 25°C with a Coefficient of 5 mV/°C. See Figure 38 for an equivalent circuit.
11 VREF General-Purpose Reference Voltage Output of 2.3 V at 25°C. See Figure 39 for an equivalent circuit.
12 VTGT The voltage applied to this pin determines the target power at the input of the RF squaring circuit. The intercept
voltage is proportional to the voltage applied to this pin. The use of a lower target voltage increases the crest
factor capacity; however, this can affect the system loop response. See Figure 44 for an equivalent circuit.
13 NC No Connect. Do not connect this pin.
14 INHI RF Input. The RF input signal is normally ac-coupled to this pin through a coupling capacitor. See Figure 37 for
an equivalent circuit.
15 INLO RF Input Common. This pin is normally ac-coupled to ground through a coupling capacitor. See Figure 37 for
an equivalent circuit.
16 NC No Connect. Do not connect this pin.
on on
Data Sheet ADL5902
Rev. B | Page 9 of 28
TYPICAL PERFORMANCE CHARACTERISTICS
VS = 5 V, ZO = 50 Ω, single-ended input drive, VOUT connected to VSET, VTGT = 0.8 V, CLPF = 0.1 µF, TA = +25°C (black), −40°C (blue),
+85°C (red), +125°C (orange) where appropriate. Error referred to the best fit line (linear regression) from − 10 dBm to − 40 dBm, unless
otherwise indicated. Input RF signal is a sine wave (CW), unless otherwise indicated.
08218-003
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
P
IN
(dBm)
T
ADJ
= 0.5V
CALIBRATION AT 0dBm, –45dBm, AND –60dBm
Figure 3. Typical VOUT and Log Conformance Error with Respect to 25°C Ideal
Line over Temperature vs. Input Amplitude at 100 MHz, CW
08218-004
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
P
IN
(dBm)
T
ADJ
= 0.4V
CALIBRATION AT 0dBm, –45dBm, AND –60dBm
Figure 4. Typical VOUT and Log Conformance Error with Respect to 25°C Ideal
Line over Temperature vs. Input Amplitude at 700 MHz, CW
08218-005
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
P
IN
(dBm)
T
ADJ
= 0.4V
CALIBRATION AT 0dBm, –45dBm, AND –60dBm
Figure 5. Typical VOUT and Log Conformance Error with Respect to 25°C Ideal
Line over Temperature vs. Input Amplitude at 900 MHz, CW
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOL
T
AGE (V)
P
IN
(dBm)
V
TADJ
= 0.5V
REPRESENTS 55 DEVICES FROM 2 LOTS
08218-006
Figure 6. Distribution of Error with Respect to 25°C over Temperature vs.
Input Amplitude, CW, Frequency = 100 MHz
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOL
T
AGE (V)
P
IN
(dBm)
V
TADJ
= 0.4V
REPRESENTS 55 DEVICES FROM 2 LOTS
08218-007
Figure 7. Distribution of Error with Respect to 25°C over Temperature vs.
Input Amplitude, CW, Frequency = 700 MHz
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
P
IN
(dBm)
V
TADJ
= 0.4V
REPRESENTS 55 DEVICES FROM 2 LOTS
08218-008
Figure 8. Distribution of Error with Respect to 25°C over Temperature vs.
Input Amplitude, CW, Frequency = 900 MHz
ADL5902 Data Sheet
Rev. B | Page 10 of 28
08218-009
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
P
IN
(dBm)
T
ADJ
= 0.4V
CALIBRATION AT 0dBm, –45dBm, AND –60dBm
Figure 9. Typical VOUT and Log Conformance Error with Respect to 25°C Ideal
Line over Temperature vs. Input Amplitude at 1.9 GHz, CW
08218-010
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
P
IN
(dBm)
T
ADJ
= 0.4V
CALIBRATION AT 0dBm, –45dBm, AND –60dBm
Figure 10. Typical VOUT and Log Conformance Error with Respect to 25°C Ideal
Line over Temperature vs. Input Amplitude at 2.14 GHz, CW
08218-011
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
P
IN
(dBm)
T
ADJ
= 0.45V
CALIBRATION AT 0dBm, –45dBm, AND –60dBm
Figure 11. Typical VOUT and Log Conformance Error with Respect to 25°C Ideal
Line over Temperature vs. Input Amplitude at 2.6 GHz, CW
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
P
IN
(dBm)
V
TADJ
= 0.4V
REPRESENTS 55 DEVICES FROM 2 LOTS
08218-012
Figure 12. Distribution of Error with Respect to 25°C over Temperature vs.
Input Amplitude, CW, Frequency = 1.9 GHz
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAG E ( V)
P
IN
(dBm)
V
TADJ
= 0.4V
REPRESENTS 55 DEVICES FROM 2 LOTS
08218-013
Figure 13. Distribution of Error with Respect to 25°C over Temperature vs.
Input Amplitude, CW, Frequency = 2.14 GHz
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
P
IN
(dBm)
V
TADJ
= 0.45V
REPRESENTS 55 DEVICES FROM 2 LOTS
0
8218-014
Figure 14. Distribution of Error with Respect to 25°C over Temperature vs.
Input Amplitude, CW, Frequency = 2.6 GHz
van
Data Sheet ADL5902
Rev. B | Page 11 of 28
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
OUTPUT VOLTAGE (V)
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
ERROR (dB)
P
IN
(dBm)
T
ADJ
= 0.5V
CALIBRATION AT 0dBm, –40dBm, AND –60dBm
08218-115
Figure 15. Typical VOUT and Log Conformance Error with Respect to 25°C Ideal
Line over Temperature vs. Input Amplitude at 3.5 GHz, CW
08218-016
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
P
IN
(dBm)
T
ADJ
= 0.95V
CALIBRATION AT 0dBm, –30dBm, AND –50dBm
Figure 16. Typical VOUT and Log Conformance Error with Respect to 25°C Ideal
Line over Temperature vs. Input Amplitude at 5.8 GHz, CW
0
100
200
300
50
150
250
350
V
OUT
(V)
COUNT
2.65 2.70 2.75 2.80 2.85 2.90 2.95 3.00 3.05
08218-017
REPRESENTS 1900
PARTS FROM 3 LOTS
Figure 17. Distribution of VOUT, PIN = −10 dBm, 900 MHz
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
PIN (dBm)
VTADJ = 0.5V
REPRESENTS 55 DEVICES FROM 2 LOTS
08218-018
Figure 18. Distribution of Error with Respect to 25°C over Temperature vs.
Input Amplitude, CW, Frequency = 3.5 GHz
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.5
1.0
1.5
2.0
2.5
3.0
–60 –50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOL
T
AGE (V)
P
IN
(dBm)
V
TADJ
= 0.95V
REPRESENTS 55 DEVICES FROM 2 LOTS
0
8218-019
Figure 19. Distribution of Error with Respect to 25°C over Temperature vs.
Input Amplitude, CW, Frequency = 5.8 GHz
0
100
200
300
50
150
250
350
V
OUT
(V)
COUNT
0.20 0.25 0.30 0.35 0.40 0.45 0.50
08218-020
REPRESENTS 1900
PARTS FROM 3 LOTS
Figure 20. Distribution of VOUT, PIN = −60 dBm, 900 MHz
:0) :0)
ADL5902 Data Sheet
Rev. B | Page 12 of 28
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
OUTPUT VOLTAGE (V)
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
ERROR (dB)
V
OUT
CW PEP = 0dB
V
OUT
QPSK PEP = 2.76
V
OUT
16 QAM PEP = 5.13
V
OUT
CDMA2000 PEP = 11.02
ERROR CW
ERROR QPSK
ERROR 16 QAM
ERROR CDMA2000
P
IN
(dBm)
08218-121
Figure 21. Error from CW Linear Reference vs. Signal Modulation,
Frequency = 900 MHz, CLPF = 0.1µF, Three-Point Calibration at 0 dBm,
−45 dBm, and −60 dBm
0
1
2
3
4
5
6
10123456789
OUTPUT VOL
T
AGE (V)
TIME (µs)
RF ENVELOPE
0dBm
–10dBm
–20dBm
–30dBm
–40dBm
08218-027
Figure 22. Output Response to RF Burst Input, Carrier Frequency 2.14 GHz,
CLPF = 220 pF, Rising Edge
0
1
2
3
4
5
6
–200 0 200 400 600 800 1000 1200 1400 1600 1800
OUTPUT VOLTAGE (V)
TIME (µs)
RF ENVELOPE
0dBm
–10dBm
–20dBm
–30dBm
–40dBm
0
8218-028
Figure 23. Output Response to RF Burst Input, Carrier Frequency 2.14 GHz,
CLPF = 0.1 F, Rising Edge
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
–60 –50 –40 –30 –20 –10 0 10
OUTPUT VOLTAGE (V)
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
ERROR (dB)
P
IN
(dBm)
V
OUT
CW PEP = 0dB
V
OUT
64 QAM PEP = 6.2dB
V
OUT
1CR W-CDMA PEP = 10.56dB
V
OUT
4CR W-CDMA
V
OUT
LTE TM1 1CR 20MHz PEP = 11.58dB
ERROR CW
ERROR 64 QAM
ERROR 1CR W-CDMA
ERROR 4CR W-CDMA
ERROR LTE TM1 1CR 20MHz
08218-124
Figure 24. Error from CW Linear Reference vs. Signal Modulation,
Frequency = 2.14 GHz, CLPF = 0.1 µF, Three-Point Calibration at −10 dBm,
−45 dBm, and −60 dBm
0
1
2
3
4
5
6
4 0 4 8 12162024283236
OUTPUT VOLTAGE (V)
TIME (µs)
RF ENVELOPE
0dBm
–10dBm
–20dBm
–30dBm
–40dBm
08218-030
Figure 25. Output Response to RF Burst Input, Carrier Frequency 2.14 GHz,
CLPF = 220 pF, Falling Edge
0
1
2
3
4
5
6
–2000 0 2000 4000 6000 8000 10,000 12,000 14,000 16,000 18,000
OUTPUT VOL
T
AGE (V)
TIME (µs)
RF ENVELOPE
0dBm
–10dBm
–20dBm
–30dBm
–40dBm
08218-031
Figure 26. Output Response to RF Burst Input, Carrier Frequency 2.14 GHz,
CLPF = 0.1 µF, Falling Edge
Data Sheet ADL5902
Rev. B | Page 13 of 28
1.29 1.32 1.35 1.38 1.41 1.44 1.47 1.50
0
100
200
300
400
V
TEMP
VOLTAGE
(V)
COUNT
08218-033
REPRESENTS 1900
PARTS FROM 3 LOTS
Figure 27. Distribution of VTEMP Voltage at 25°C, No RF Input
0
100
200
300
400
V
REF
BIAS VOLTAGE (V)
COUNT
2.19 2.22 2.25 2.28 2.31 2.34 2.37 2.40 2.43
08218-034
REPRESENTS 1900
PARTS FROM 3 LOTS
Figure 28. Distribution of VREF Voltage at 25°C, No RF Input
–40
–30
–20
–10
0
10
20
30
40
–55 –35 –15 5 25 45 65 85 105 125
TEMPERATUREC)
CHANGE IN V
REF
(mV)
08218-037
Figure 29. Change in VREF vs. Temperature with Respect to 25°C,
RF Input = −40 dBm, Typical Device
0.5
0.7
0.9
1.1
1.3
1.5
1.7
1.9
2.1
2.3
2.5
–55 –35 –15 5 25 45 65 85 105 125
TEMPERATURE (°C)
–2.5
–2.0
–1.5
–1.0
–0.5
0
0.5
1.0
1.5
2.0
2.5
ERROR (°C)
V
TEMP
(V)
08218-036
Figure 30. VTEMP and Linearity Error with Respect to Straight Line vs.
Temperature for Typical Device
–1.0
–0.8
–0.6
–0.4
–0.2
0
0.2
–50 –40 –30 –20 –10 0 10 20
CHANGE IN V
REF
(mV)
P
IN
(dBm)
08218-035
Figure 31. Change in VREF vs. Input Amplitude with Respect to −40 dBm,
25°C, Typical Device
0.1
1
10
100
4.0 4.1 4.2 4.3 4.4 4.5 4.6 4.7 4.8 4.9 5.0
SUPPLY CURRENT (mA)
V
PWDN
(V)
V
PWDN
INCREASING
V
PWDN
DECREASING
08218-038
Figure 32. Supply Current vs. VPWDN
.maa \ .3043". H V / \ \ /
ADL5902 Data Sheet
Rev. B | Page 14 of 28
0
1
2
3
4
5
6
7
–4 0 4 8 12 16 20 24 28 32
OUTPUT VOLTAGE (V)
TIME (µs)
TADJ/PWDN PULSE
0dBm
–10dBm
–20dBm
–30dBm
–40dBm
08218-032
Figure 33. Output Response Using Power-Down Mode for Various RF Input
Levels Carrier Frequency 2.14 GHz, CLPF = 220 pF
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
0123456789
FREQUENCY (GHz)
–10dBm
–30dBm
V
OUT
(V)
0
8218-026
Figure 34. Typical VOUT vs. Frequency for Two RF Input Amplitudes,
50 MHz to 9 GHz
0
20
40
60
80
100
120
140
160
180
200
100 1k 10k 100k 1M 10M
NOISE SPECTR
A
L DENSI
T
Y (nV/
Hz)
FREQUENCY (Hz)
08218-039
Figure 35. Noise Spectral Density of VOUT, RF Input = −20 dBm, All CLPF Values
Data Sheet ADL5902
Rev. B | Page 15 of 28
THEORY OF OPERATION
The ADL5902 is a 50 MHz to 9 GHz true rms responding
detector with a 65 dB measurement range at 2.14 GHz and a
greater than 56 dB measurement range at frequencies up to 6 GHz.
It incorporates a modified AD8362 architecture that increases the
frequency range and improves measurement accuracy at high
frequencies. Transfer function peak-to-peak ripple is reduced to
<±0.1 dB over the entire dynamic range. Temperature stability of
the rms output measurements provides <±0.3 dB error, typically,
over the specified temperature range of −4C to 125°C through
proprietary techniques. The device accurately measures waveforms
that have a high peak-to-rms ratio (crest factor).
The ADL5902 consists of a high performance AGC loop. As
shown in Figure 36, the AGC loop comprises a wide bandwidth
variable gain amplifier (VGA), square law detectors, an amplitude
target circuit, and an output driver. For a more detailed description
of the functional blocks, see the AD8362 data sheet.
The nomenclature used in this data sheet to distinguish
between a pin name and the signal on that pin is as follows:
The pin name is all uppercase, for example, VPOS,
COMM, and VOUT.
The signal name or a value associated with that pin is the
pin mnemonic with a partial subscript, for example, CLPF
and VOUT.
SQUARE LAW DETECTOR AND AMPLITUDE TARGET
The VGA gain has the form
GSET = GO e )/( GNSSET VV (1)
where:
GO is the basic fixed gain.
VGNS is a scaling voltage that defines the gain slope (the decibel
change per voltage). The gain decreases with increasing VSET.
The VGA output is
VSIG = GSET × RFIN = GO × RFIN e )/( GNSSET VV (2)
where RFIN is the ac voltage applied to the input terminals of the
ADL5902.
The output of the VGA, VSIG, is applied to a wideband square
law detector. The detector provides the true rms response of the
RF input signal, independent of waveform. The detector output,
ISQR, is a fluctuating current with positive mean value. The
difference between ISQR and an internally generated current, ITGT, is
integrated by the parallel combination of CF and the external
capacitor attached to the CLPF pin at the summing node. CF is
an on-chip 26 pF filter capacitor, and CLPF, the external capacitance
connected to the CLPF pin, can arbitrarily increase the averaging
time while trading off with the response time. When the AGC
loop is at equilibrium
Mean(ISQR) = ITGT (3)
This equilibrium occurs only when
Mean(VSIG2) = VTGT2 (4)
where VTGT is the voltage presented at the VTGT pin. This pin
can conveniently be connected to the VREF pin through a voltage
divider to establish a target rms voltage, VATG, of ~40 mV rms when
VTGT = 0.8 V.
Because the square law detectors are electrically identical and
well matched, process and temperature dependent variations
are effectively cancelled.
TADJ/PWDN
BAND GAP
REFERENCE
COMM
VOUT
TEMP (1.4V)
VREF (2.3V)
I
SQR
I
TGT
X
2
X
2
G
SET
C
LPF
(EXTERNAL)
C
F
(INTERNAL)
V
SIG
VGA
SUMMING
NODE
V
SET
C
H
(INTERNAL)
V
POS
INHI
INLO
TEMPERATURE COMPENSATION
AND BIAS
TEMPERATURE
SENSOR
VTGT
CLPF
V
ATG
= V
TGT
20
08218-040
Figure 36. Simplified Architecture Details
1i
ADL5902 Data Sheet
Rev. B | Page 16 of 28
When forcing the previous identity by varying the VGA setpoint, it
is apparent that
RMS(VSIG) = √(Mean(VSIG2)) = √(VATG2) = VATG (5)
Substituting the value of VSIG from Equation 2 results in
RMS(G0 × RFIN e )/( GNSSET VV) = VATG (6)
When connected as a measurement device, VSET = VOUT. Solving
for VOUT as a function of RFIN,
VOUT = VSLOPE × log10(RMS(RFIN)/VZ) (7)
where:
VSLOPE is 1.06 V/decade (or 53 mV/dB) at 2.14 GHz.
VZ is the intercept voltage.
When RMS(RFIN) = VZ, this implies that VOUT = 0 V because
log10(1) = 0. This makes the intercept the input that forces VOUT =
0 V if the ADL5902 had no sensitivity limit. The PINTERCEPT (in
decibels relative to 1 milliwatt, that is, dBm) corresponding to
Vz (in volts) in ADL5902 is given by the following equation:
PINTERCEPT = −(VPEDISTAL/VSLOPE) + PMINDET (8)
where VPEDISTAL is the VSET interface pedestal voltage, and
PMINDET is the minimum detectable signal in decibels relative to
1 milliwatt, given by the following expression:
PMINDET = dBm (VATG) – GO (9)
where dBm(VATG) is the equivalent power in decibels relative to
1 milliwatt corresponding to a given VTGT.
Combining Equation 8 and Equation 9 results in
PINTERCEPT = −(VPEDISTAL/VSLOPE) + dBm (VATG) – GO (10)
For the ADL5902, VPEDISTAL is approximately 0.275 V and VATG is
given by VTGT/20. GO is 45 dB below approximately 4 GHz and
then decreases at higher frequencies. VTGT = 0.8 V; therefore,
VATG = 40 mV
and
dBm (VATG) = 10 log10((40 mV)2/50 Ω)/1 mW) ≈ −14.9 dBm
At 2.14 GHz, VSLOPE ≈ 53 mV/dB and GO at 2.14 GHz = 45 dB.
This results in a PINTERCEPT ≈ −65 dBm. This differs slightly from
the value in Table 1 due to the choice of calibration points and
the slight nonideality of the response.
In most applications, the AGC loop is closed through the setpoint
interface and the VSET pin. In measurement mode, VOUT is
directly connected to VSET (see the Measurement Mode Basic
Connections section for more information). In controller mode, a
control voltage is applied to VSET, and the VOUT pin typically
drives the control input of an amplification or attenuation system.
In this case, the voltage at the VSET pin forces a signal amplitude
at the RF inputs of the ADL5902 that balances the system
through feedback.
RF INPUT INTERFACE
Figure 37 shows the RF input connections within the ADL5902.
The input impedance is set primarily by an internal 2 kΩ resistor
connected between INHI and INLO. A dc level of approximately
half the supply voltage on each pin is established internally. Either
the INHI or INLO pin can be used as the single-ended RF input
pin. Signal coupling capacitors must be connected from the input
signal to the INHI and INLO pins. A single external 60.4 Ω resistor
to ground from the desired input creates an equivalent 50 Ω
impedance over a broad section of the operating frequency range.
The other input pin must be RF ac-coupled to common (ground).
The input signal high-pass corner formed by the input coupling
capacitor internal and external resistances is
fHIGHPASS = 1/(2 × π × 50 × C) (11)
where C is the capacitance in farads and fHIGHPASS is in hertz. The
input coupling capacitors must be large enough in value to pass
the input signal frequency of interest and determine the low end
of the frequency response. INHI and INLO can also be driven
differentially using a balun.
ESD
ESD
ESD ESD ESD ESD
ESD
ESD ESD ESD ESD ESD
ESD
INLOINHI
VPOS
COMM
BIAS
LOAD
2k2k
08218-041
Figure 37. RF Inputs
Extensive ESD protection is employed on the RF inputs, and
this protection limits the maximum possible input to the ADL5902.
Data Sheet ADL5902
Rev. B | Page 17 of 28
SMALL SIGNAL LOOP RESPONSE
The ADL5902 uses a VGA in a loop to force a squared RF signal
to be equal to a squared dc voltage. This nonlinear loop can be
simplified and solved for a small signal loop response. The low-
pass corner pole is given by
FreqLP ≈ 1.83 × ITGT/(CLPF) (12)
where:
ITGT is in amperes.
CLPF is in farads.
FreqLP is in hertz.
ITGT is derived from VTGT; however, ITGT is a squared value of
VTGT multiplied by a transresistance, namely
ITGT = gm × VTGT2 (13)
gm is approximately 18.9 µs; therefore, with VTGT equal to the
typically recommended 0.8 V, ITGT is approximately 12 µA. The
value of this current varies with temperature; therefore, the small
signal pole varies with temperature. However, because the RF
squaring circuit and dc squaring circuit track with temperature,
there is no temperature variation contribution to the absolute value
of VOUT.
For CW signals,
FreqLP ≈ 67.7 × 10−6/(CLPF) (14)
However, signals with large crest factors include low pseudo-
random frequency content that must be either filtered out or
sampled and averaged out (see the Choosing a Value for CLPF
section for more information).
TEMPERATURE SENSOR INTERFACE
The ADL5902 provides a temperature sensor output with a scaling
factor of the output voltage of approximately 4.9 mV/°C. The
output is capable of sourcing 4 mA and sinking 50 A maximum
at 2C. An external resistor can be connected from TEMP to
COMM to provide additional current sink capability. The
typical output voltage at 25°C is approximately 1.4 V.
08218-042
TEMP
V
POS
COMM
INTERNAL
VPAT
12k
4k
Figure 38. TEMP Interface Simplified Schematic
VREF INTERFACE
The VREF pin provides an internally generated voltage reference
for the user. The VREF voltage is a temperature stable 2.3 V
reference that is capable of sourcing 4 mA and sinking 50 A
maximum. An external resistor can be connected from VREF to
COMM to provide additional current sink capability. The voltage
on this pin can drive the TADJ/PWDN and VTGT pins.
08218-143
INTERNAL
VOLTAGE
16k
VREF
V
POS
COMM
Figure 39. VREF Interface Simplified Schematic
TEMPERATURE COMPENSATION INTERFACE
While the ADL5902 has a highly stable measurement output
with respect to temperature using proprietary techniques, for
optimal performance, the output temperature drift must be
compensated for using the TADJ pin. The absolute value of
compensation varies with frequency and VTGT. Table 4 shows the
recommended voltages for VTADJ to maintain a temperature drift
error of typically ±0.5 dB or better over the intended temperature
range (−40°C < TA < +85°C) when driven single-ended and
VTGT = 0.8 V.
Table 4. Recommended VTADJ for Selected Frequencies
Frequency VTADJ (V) R9 in Figure 54 (Ω) R12 in Figure 54 (Ω)
100 MHz 0.5 1430 402
700 MHz 0.4 1430 301
900 MHz 0.4 1430 301
1.9 GHz 0.4 1430 301
2.14 GHz 0.4 1430 301
2.6 GHz 0.45 1430 348
3.5 GHz 0.5 1430 402
5.8 GHz 0.95 1430 1007
The values in Table 4 are chosen to give the best drift
performance at the high end of the usable dynamic range
over the −40°C to +85°C temperature range. There is often a
trade off in setting values, and optimizing for one area of the
dynamic range can mean less than optimal drift performance at
other input amplitudes.
// I/ \ /l //I\ \ \ \ 4m:
ADL5902 Data Sheet
Rev. B | Page 18 of 28
Compensating the device for temperature drift using TADJ allows
for great flexibility. If the user requires minimum temperature
drift at a given input power, a subset of the dynamic range, or
even over a different temperature range than shown in this data
sheet, the VTADJ can be swept while monitoring VOUT over the
temperature at the frequency and amplitude of interest. The
optimal VTADJ to achieve minimum temperature drift at a given
power and frequency is the value of VTADJ where the output has
minimum movement.
2.73
2.75
2.77
2.79
2.81
2.83
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8
+125°C
+105°C
+85°C
+55°C
+25°C
–2C
–40°C
C
V
OUT
(V)
V
TADJ
(V)
08218-044
Figure 40. Effect of VTADJ at Various Temperatures, 2.14 GHz, −10 dBm
Var ying VTADJ has only a very slight effect on VOUT at device
temperatures near 25°C; however, the compensation circuit
has more and more effect as the temperature departs farther
from 25°C.
The TADJ pin has a high input impedance and can be conven-
iently driven from an external source or from an attenuated value
of VREF using a resistor divider. Table 4 gives suggested voltage
divider values to generate the required voltage from VREF. The
resistors are shown in the evaluation board schematic (see
Figure 54). VREF does change slightly with temperature and also
input RF amplitude; however, the amount of change is unlikely
to result in a significant effect on the final temperature stability
of the RF measurement system. Typically, the temperature comp-
ensation circuit responds only to voltages between 0 and VS/2,
or about 2.5 V when VS = 5 V.
Figure 41 in the Power-Down Interface section shows a simpli-
fied schematic representation of the TADJ/PWDN interface.
POWER-DOWN INTERFACE
The quiescent and disabled currents for the ADL5902 at 25°C
are approximately 73 mA and 300 µA, respectively. The dual
function TADJ/PWDN pin is connected to the temperature comp-
ensation circuit as well as the power-down circuit. Typically, the
temperature compensation circuit responds only to voltages
between 0 and VS/2, or about 2.5 V when VS = 5 V.
When the voltage on this pin is greater than VS − 0.1 V, the device
is fully powered down. Figure 32 shows this characteristic as a
function of VPWDN. Note that, because of the design of this section
of the ADL5902, as VPWDN passes through a narrow range at
~4.5 V (or ~VS − 0.5 V), the TADJ/PWDN pin sinks approximately
500 µA. The source used to disable the ADL5902 must have a
sufficiently high current capability for this reason. Figure 33
shows the typical response times for various RF input levels.
The output reaches within 0.1 dB of the steady-state value in
approximately 5 µs; however, the reference voltage is available to
full accuracy in a much shorter time. This wake-up response
varies depending on the input coupling and CLPF.
08218-076
TADJ/
PWDN
COMM
V
POS
200
200
7k7k
VREF
INTERCEPT
TEMPERATURE
COMPENSATION
200
POWER-UP
CIRCUIT
SHUTDOWN
CIRCUIT
ESD
ESD
ESD
Figure 41. TADJ/PWDN Interface Simplified Schematic
VSET INTERFACE
The VSET interface has a high input impedance of 72 kΩ. The
voltage at VSET is converted to an internal current used to set
the internal VGA gain. The VGA attenuation control is approx-
imately 19 dB/V.
08218-149
ACOM
2.5k
18k
VSET
GAIN ADJUST
54k
Figure 42. VSET Interface Simplified Schematic
OUTPUT INTERFACE
The ADL5902 incorporates rail-to-rail output drivers with pull-
up and pull-down capabilities. The closed-loop, − 3dB bandwidth
from the input of the output amplifier to the output with no load is
approximately 58 MHz with a single-pole roll off of approximately
−20 dB/decade. The output noise is approximately 25 nV/√Hz
at 100 kHz. The VOUT pin can source and sink up to 10 mA.
There is also an internal load from VOUT to COMM of 2500 Ω.
08218-045
VOUT
CLPF
2k
500
2pF
ESD
ESD
ESD
V
POS
COMM
Figure 43. VOUT Interface Simplified Schematic
Data Sheet ADL5902
Rev. B | Page 19 of 28
VTGT INTERFACE
The target voltage can be set with an external source or by
connecting the VREF pin (nominally 2.3 V) to the VTGT pin
through a resistive voltage divider. With 0.8 V on the VTGT pin,
the rms voltage that must be provided by the VGA to balance the
AGC feedback loop is 0.8 V × 0.05 = 40 mV rms. Most of the
characterization information in this data sheet was collected at
VTGT = 0.8 V. Voltages higher and lower than this can be used;
however, doing so increases or decreases the gain at the internal
squaring cell, which results in a corresponding increase or decrease
in intercept. This, in turn, affects the sensitivity and the usable
measurement range, in addition to the sensitivity to different
carrier modulation schemes. As VTGT decreases, the squaring
circuits produce more noise; this becomes noticeable in the output
response at low input signal amplitudes. As VTGT increases,
measurement error due to modulation increases and temperature
drift tends to decrease. The chosen VTGT value of 0.8 V represents a
compromise between these characteristics.
08218-048
VTGT 50k
50k
10k
ESD
ESD
ESD
V
POS
COMM
g × X
2
ITGT
Figure 44. VTGT Interface
BASIS FOR ERROR CALCULATIONS
The slope and intercept used in the error plots are calculated using
the coefficients of a linear regression performed on data collected
in the central operating range. The error plots in the Typical
Performance Characteristics section are shown in two formats:
error from the ideal line and error with respect to the 25°C output
voltage. The error from the ideal line is the decibel difference in
VOUT from the ideal straight-line fit of VOUT calculated by the
linear-regression fit over the linear range of the detector, typically
at 25°C. The error in decibels is calculated by
Error (dB) = (VOUTSlope × (PINPZ))/Slope (15)
where PZ is the x-axis intercept expressed in decibels relative to
1 milliwatt (the input amplitude produces a 0 V output if such an
output is possible).
The error from the ideal line is not a measure of absolute accuracy
because it is calculated using the slope and intercept of each device.
However, it verifies the linearity and the effect of temperature
and modulation on the response of the device.
An example of this type of plot is Figure 3. The slope and
intercept that form the ideal line are those at 25°C with CW
modulation. Figure 21 and Figure 24 show the error with various
popular forms of modulation with respect to the ideal CW line.
This method for calculating error is accurate, assuming that
each device is calibrated at room temperature.
In the second plot format, the VOUT voltage at a given input
amplitude and temperature is subtracted from the corresponding
VOUT at 25°C and then divided by the 25°C slope to obtain an error
in decibels. This type of plot does not provide any information
on the linear-in-dB performance of the device; it merely shows
the decibel equivalent of the deviation of VOUT over temperature,
given a calibration at 2C. When calculating error from any
one particular calibration point, this error format is accurate. It
is accurate over the full range shown on the plot assuming that
enough calibration points are used. Figure 6 shows this plot type.
The error calculations for Figure 30 are similar to those for the
VOUT plots. The slope and intercept of the VTEMP function vs.
temperature are determined and applied as follows:
Error (°C) = (VTEMPSlope × (TempTZ))/Slope (16)
where:
TZ is the x-axis intercept expressed in degrees Celsius (the temp-
erature that results in a VTEMP of 0 V if possible).
Temp is the ambient temperature of the ADL5902 in degrees
Celsius.
Slope is, typically, 4.9 mV/°C.
VTEMP is the voltage at the TEMP pin at that temperature.
MEASUREMENT MODE BASIC CONNECTIONS
Figure 45 shows the basic connections for operating the ADL5902
as they are implemented on the device evaluation board. The
ADL5902 requires a single supply of nominally 5 V. The supply
is connected to the two VPOS supply pins. These pins must
each be decoupled using the two capacitors with values equal or
similar to those shown in Figure 45. These capacitors must be
placed as close as possible to the VPOS pins.
An external 60.4  resistor (R3) combines with the relatively high
RF input impedance of the ADL5902 to provide a broadband 50 
match. An ac coupling capacitor must be placed between this
resistor and INHI. The INLO input must be ac-coupled to ground
using the same value capacitor. Because the ADL5902 has a
minimum input operating frequency of 50 MHz, 100 pF ac
coupling capacitors can be used.
The ADL5902 is placed in measurement mode by connecting
VOUT to VSET. In measurement mode, the output voltage is
proportional to the log of the rms input signal level.
ADL5902 Data Sheet
Rev. B | Page 20 of 28
08218-145
X
2
X
2
BIAS AND POWER-
DOWN CONTROL
1
NC
NC
NC
LINEAR-IN-dB VGA
(NEGATIVE SLOPE)
I
DET
26pF
2
3
4
11
10
9
5
6
7
8
16
15
14
13
ADL5902
12
VREF
2.3V
TEMPERATURE
SENSOR
INLO
INHI
VPOS POS
TEMP
(BLACK)
GND
(BLACK)
VSET
(BLACK)
VOUT
(BLACK)
VTGT
(BLACK)
VREF
(BLACK)
TEMP
VSET
VOUT
CLPF
COMM
COMMVTGTVREFTADJ/PWDN
G = 5
I
TGT
R3
60.4
C10
100pF
C12
100pF
V
POS
+5V
(RED)
C4
100pF
C3
0.1µF
C5
100pF
C7
0.1µF
R11
2k
R10
3.74k
R12
301
R9
1430
C9
10µF
RFIN R2
OPEN
R6
0
R1
0
R15
OPEN
TC2 PWDN
(BLACK)
Figure 45. Basic Connections for Operation in Measurement Mode
SETTING VTADJ
As discussed in the Theory of Operation section, the output
temperature drift must be compensated by applying a voltage to
the TADJ pin. The compensating voltage varies with frequency.
The voltage for the TADJ pin can be easily derived from a resistor
divider connected to the VREF pin. Table 5 shows the recom-
mended VTADJ for operation from −40°C to +85°C, along with
resistor divider values. Resistor values are chosen so that they
neither pull too much current from VREF (VREF short-circuit
current is 4 mA) nor are so large that the TADJ pin bias current of
3 µA affects the resulting voltage at the TADJ pin.
Table 5. Recommended VTADJ for Selected Frequencies
Frequency VTA DJ (V) R9 (Ω) R12 (Ω)
100 MHz 0.5 1430 402
700 MHz to 2.14 GHz 0.4 1430 301
2.6 GHz 0.45 1430 348
3.5 GHz 0.5 1430 402
5.8 GHz 0.95 1430 1007
SETTING VTGT
As discussed in the Theory of Operation section, setting the
voltage on VTGT to 0.8 V represents a compromise between
achieving excellent rms compliance and maximizing dynamic
range. The voltage on VTGT can be derived from the VREF pin
using a resistor divider as shown Figure 45 (Resistor R10 and
Resistor R11). Like the resistors chosen to set the VTADJ voltage,
the resistors setting VTGT must have reasonable values that do
not pull too much current from VREF or cause bias current
errors. Also, attention must be paid to the combined current
that VREF must deliver to generate the VTADJ and VTGT voltages.
This current must be kept well below the VREF short-circuit
current of 4 mA.
CHOOSING A VALUE FOR CLPF
CLPF (C9 in Figure 45) provides the averaging function for the
internal rms computation. Using the minimum value for CLPF
allows the quickest response time to a pulsed waveform but
leaves significant output noise on the output voltage signal. By
the same token, a large filter cap reduces output noise but at the
expense of response time.
For non response-time critical applications, a relatively large
capacitor can be placed on the CLPF pin. In Figure 45, a value
of 0.1 µF is used. For most signal modulation schemes, this
value ensures excellent rms measurement compliance and low
residual output noise. There is no maximum capacitance limit
for CLPF.
\ \ Y\
Data Sheet ADL5902
Rev. B | Page 21 of 28
Figure 46 shows how output noise varies with CLPF when the
ADL5902 is driven by a single-carrier W-CDMA signal (Test
Model TM1-64, peak envelope power = 10.56 dB, bandwidth =
3.84 MHz). With a 10 µF capacitor on CLPF, there is residual
noise on VOUT of 4.4 mV p-p, which is less than 0.1 dB error
(assuming a slope of approximately 53 mV/dB).
08218-146
RISE/FALL TIME (µs)
OUTPUT NOISE (mV p-p)
1
10
100
1k
10k
100k
1M
0
50
100
150
200
250
300
1 10 100 1000
C
LPF
(nF)
OUTPUT NOISE (mV p-p)
10% TO 90% RISE TIME (µs)
90% TO 10% FALL TIME (µs)
Figure 46. Output Noise, Rise and Fall Times vs. CLPF Capacitance, Single-
Carrier W-CDMA (TM1-64) at 2.14 GHz with PIN = 0 dBm
Figure 46 also shows how CLPF affects the response time of VOUT.
To measure this, a RF burst at 2.14 GHz at −10 dBm was applied to
the ADL5902. The 10% to 90% rise time and 90% to 10% fall
time is then measured. It is notable that the fall time is much
longer than the rise time. This can also be seen in the response
time plots, Figure 22, Figure 23, Figure 25, and Figure 26.
In applications where the response time is critical, a different
approach to signal filtering can be taken. This is shown in
Figure 47. The capacitor on the CLPF pin is set to the minimum
value that ensures that a valid rms computation is performed.
The job of noise removal is then handed off to an RC filter on
the VOUT pin. This approach ensures that there is enough
averaging to ensure good rms compliance and does not burden
the rms computation loop with extra filtering that significantly
slows down the response time. By finishing the filtering process
using an RC filter after VOUT, faster fall times can be achieved
with an equivalent amount of output noise. It must be noted
that the RC filter can also be implemented in the digital domain
after the analog-to-digital converter.
In Figure 47, CLPF is equal to 10 nF. This value was experimentally
determined to be the minimum capacitance that ensures good
rms compliance when the ADL5902 is driven by a 1 C W-CDMA
signal (TM1-64). This test was carried out by starting out with a
large capacitance value on the CLPF pin (for example, 10 µF).
The value of VOUT was noted for a fixed input power level (for
example, −10 dBm). The value of CLPF was then progressively
reduced (this can be done with press-down capacitors) until
the value of VOUT started to deviate from the original value (this
indicates that the accuracy of the rms computation is degrading
and that CLPF is getting too small).
0
8218-147
X
2
BIAS AND POWER-
DOWN CONTROL
1
NC
NC
NC
LINEAR-IN-dB VGA
(NEGATIVE SLOPE)
I
DET
26pF
2
3
4
11
10
9
5
6
7
8
16
15
14
13
ADL5902
12
VREF
2.3V
TEMPERATURE
SENSOR
INL
O
INHI
V
POS POS
TEMP
VSET
VOUT
CLPF
COMMCOMMVTGTVREFTADJ/PWDN
G = 5
I
TGT
X
2
C9
10nF
(SEE TABLE 6 AND
FIGURE 46.)
C
FILTER
(SEE FIGURE 48.)
R
FILTER
2k
VOUT
Figure 47. Optimizing Setting Time and Residual Ripple
mm
ADL5902 Data Sheet
Rev. B | Page 22 of 28
Figure 48 shows the resulting rise and fall times (signal is pulsed
between off and −10 dBm) with CLPF equal to 10 nF. A 2 kΩ
resistor is placed in series with the VOUT pin, and the capacitance
from this resistor to ground (CFILTER in Figure 47) is varied
up to 1 µF.
08218-148
1
10
100
1k
10k
100k
1M
0
50
100
150
200
250
300
1 10 100 1k
RISE/FALL TIME (µs)
RESIDU
A
L RIPPLE (mV p-p)
C
FILTER
(nF)
RESIDUAL RIPPLE (V p-p)
10% TO 90% RISE TIME (µs)
90% TO 10% FALL TIME (µs)
Figure 48. Residual Ripple, Rise and Fall Times Using an RC Low-Pass Filter
at VOUT, PIN = 0 dBm at 2.14 GHz
For large values of CFILTER, the fall time is dramatically reduced
compared to Figure 46. This comes at the expense of a moderate
increase in rise time.
As CFILTER is reduced, the fall time flattens out. This is because
the fall time is now dominated by the 10 nF CLPF which is
present throughout the measurement.
Table 6 shows recommended minimum values of CLPF for
popular modulation schemes, using just a single filter capacitor
at the CLPF pin. Using lower capacitor values results in rms
measurement errors. Output response time (10% to 90%) is also
shown. If the output noise shown in Table 6 is unacceptably
high, it can be reduced by
Increasing CLPF
Adding an RC filter at VOUT, as shown in Figure 47
Implementing an averaging algorithm after the ADL5902
output voltage is digitized by an ADC
Table 6. Recommended Minimum CLPF Values for Various Modulation Schemes
Modulation/Standard
Peak-Envelope
Power
Signal
Bandwidth CLPF (min) Output Noise Rise/Fall Time (10% to 90%)
W-CDMA, One-Carrier, TM1-64 10.56 dB 3.84 MHz 10 nF 95 mV p-p 12/330 s
W-CDMA Four-Carrier, TM1-64, TM1-32,
TM1-16, TM1-8
12.08 dB 18.84 MHz 5.6 nF 164 mV p-p 7/200 s
LTE, TM1 1CR 20 MHz (2048 Subcarriers,
QPSK Subcarrier Modulation)
11.58 dB 20 MHz 1000 pF 452 mV p-p 1.3/38 s
Data Sheet ADL5902
Rev. B | Page 23 of 28
OUTPUT VOLTAGE SCALING
The output voltage range of the ADL5902 (nominally 0.3 V to
3.5 V) can be easily increased or decreased. There are a number
of situations where adjustment of the output scaling makes sense.
For example, if the ADL5902 is driving an analog-to-digital
converter (ADC) with a 0 V to 5 V input range, it makes sense
to increase the detector nominal maximum output voltage of
3.5 V so that it is closer to 5 V. This makes better use of the input
range of the ADC and maximizes the resolution of the system
in terms of bits/dB. For more information on interfacing the
ADL5902 to an ADC, please refer to Circuit Note CN0178.
If only a part of the ADL5902 RF input power range is being
used (for example, −10 dBm to −60 dBm), it can make sense to
increase the scaling so that this reduced input range fits into the
ADL5902 available output swing of 0 V to 4.8 V.
The output swing can also be reduced by simply adding a
voltage divider on the output pin, as shown in the circuit on the
left-hand side of Figure 49. Reducing the output scaling can, for
example, be used when interfacing the ADL5902 to an ADC
with a 0 V to 2.5 V input range. Recommended scaling resistors
for a slope decrease are provided in Table 7.
The output voltage swing can be increased using a technique
that is analogous to setting the gain of an op amp in noninverting
mode with the VSET pin being the equivalent of the inverting
input of the op amp. This is shown in the circuit on the left-hand
side of Figure 49.
Connecting VOUT to VSET results in the nominal 0 V to 3.5 V
swing and a slope of approximately 53 mV/dB (this varies slightly
with frequency). Figure 49 and Table 7 show the configurations
for increasing the slope, along with recommended standard
resistor values for particular input ranges and output swings.
6
7
VSET
R6
R2
VOUT
6
7
VSET
08218-049
R1
R15
VOUT
Figure 49. Decreasing and Increasing Slope
Table 7. Output Voltage Range Scaling
Desired
Input
Range
(dBm)
R6
(Ω)
R2
(Ω)
R1
(Ω)
R15
(Ω)
New
Slope
(mV/dB)
Nominal
Output
Voltage
Range (V)
0 to −60 665 2000 72.1 0.195 to 4.52
−10 to −50 1180 2000 86.3 1.096 to 4.55
0 to −60 806 2000 38.3 0.103 to 2.49
−10 to −50 324 2000 46.2 0.587 to 2.43
Equation 17 is the general function that governs this.
1)||(6 '
O
O
IN V
V
RR2R
(17)
where:
VO is the nominal maximum output voltage (see Figure 6
through Figure 18).
V'O is the new maximum output voltage (for example, up to 4.8 V).
RIN is the VSET input resistance (72 kΩ).
When choosing R6 and R2, attention must be paid to the current
drive capability of the VOUT pin and the input resistance of the
VSET pin. The choice of resistors must not result in excessive
current draw out of VOUT. However, making R6 and R2 too
large is also problematic. If the value of R2 is compatible with the
input resistance of the VSET input (72 kΩ), this input resistance,
which varies slightly from device to device, contributes to the
resulting slope and output voltage. In general, the value of R2
must be at least ten times smaller than the input resistance of
VSET. Values for R6 and R2 must, therefore, be in the 1 k to
5 k range.
It is also important to take into account device to device and
frequency variation in output swing along with the ADL5902
output stage maximum output voltage of 4.8 V. The VOUT
distribution is well characterized at major frequencies’ bands in
the Typical Performance Characteristics section (see Figure 6
through Figure 8, Figure 12 through Figure 14, Figure 18, and
Figure 19). The resistor values in Table 7, which are calculated
based on 900 MHz performance, are conservatively chosen so
that there is no chance that the output voltages exceed the
ADL5902 output swing or the input range of a 0 V to 2.5 V and
0 V to 5 V ADC. Because the output swing does not vary much
with frequency (it does start to drop off above 3 GHz), these
values work for multiple frequencies.
ADL5902 Data Sheet
Rev. B | Page 24 of 28
SYSTEM CALIBRATION AND ERROR CALCULATION
The measured transfer function of the ADL5902 at 2.14 GHz is
shown in Figure 50, which contains plots of both output voltage
vs. input amplitude (power) and calculated error vs. input level. As
the input level varies from −62 dBm to +3 dBm, the output
voltage varies from ~0.25 V to ~3.5 V.
08218-050
0
1
2
3
4
5
6
–70 –60 –50 –40 –30 –20 –10 100
V
OUT
ERROR 2-POINT CAL AT 0dBm, AND 40dBm
ERROR 3-POINT CAL AT 0 dBm,
–45dBm, AND 60dBm
ERROR 4-POINT CAL AT 0dBm, –20dBm,
–45dBm, AND –60dBm
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
ERROR (dB)
V
OUT
(V)
P
IN
(dBm)
Figure 50. 2.14 GHz Transfer Function, Using Various Calibration Techniques
Because slope and intercept vary from device to device, board-
level calibration must be performed to achieve high accuracy.
The equation for the idealized output voltage can be written as
VOUT(IDEAL) = Slope × (PINIntercept) (18)
where:
Slope is the change in output voltage divided by the change in
input power (dB).
Intercept is the calculated input power level at which the output
voltage is 0 V (note that Intercept is an extrapolated theoretical
value not a measured value).
In general, calibration is performed during equipment manu-
facture by applying two or more known signal levels to the
input of the ADL5902 and measuring the corresponding output
voltages. The calibration points are generally within the linear-
in-dB operating range of the device.
With a two-point calibration, the slope and intercept are
calculated as follows:
Slope = (VOUT1VOUT2)/(PIN1PIN2) (19)
Intercept = PIN1 − (VOUT1/Slope) (20)
After the slope and intercept are calculated and stored in non-
volatile memory during equipment calibration, an equation can
calculate an unknown input power based on the output voltage
of the detector.
PIN (Unknown) = (VOUT1(MEASURED)/Slope) + Intercept (21)
The log conformance error is the difference between this
straight line and the actual performance of the detector.
Error (dB) = (VOUT(MEASURED)VOUT(IDEAL))/Slope (22)
Figure 50 includes a plot of this error when using a two-point
calibration (calibration points are 0 dBm and −40 dBm). The
error at the calibration points (in this case, −40 dBm and 0 dBm)
is equal to 0 by definition.
The residual nonlinearity of the transfer function that is
apparent in the two-point calibration error plot can be reduced
by increasing the number of calibration points. Figure 50 shows
the postcalibration error plots for three-point and four-point
calibrations. With a multipoint calibration, the transfer function
is segmented, with each segment having a slope and intercept.
Multiple known power levels are applied, and multiple voltages are
measured. When the equipment is in operation, the measured
voltage from the detector first determines which of the stored
slope and intercept calibration coefficients are to be used. Then
the unknown power level is calculated by inserting the
appropriate slope and intercept into Equation 21.
Figure 51 shows the output voltage and error at 25°C and over
temperature when a four-point calibration is used (calibration
points are 0 dBm, −20 dBm, −45 dBm, and −60 dBm). When
choosing calibration points, there is no requirement for, or
value, in equal spacing between the points. There is also no
limit to the number of calibration points used. However, using
more calibration points increases calibration time.
0
1
2
3
4
5
6
V
OUT
(V)
08218-051
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
ERROR (dB)
–70 –60 –50 –40 –30 –20 –10 100
P
IN
(dBm)
+85°C V
OUT
+25°C V
OUT
–40°C V
OUT
+85°C ERROR 4-POINT CAL
+25°C ERROR 4-POINT CAL AT 0dBm,
–20dBm, –45dBm, AND –60dBm
–40°C ERROR 4-POINT CAL
Figure 51. 2.14 GHz Transfer Function and Error at +25°C, −40°C, and +85°C
Using a Four-Point Calibration (0 dBm, −20 dBm, −45 dBm, −60 dBm)
The −40°C and +85°C error plots in Figure 51 are generated
using the 25°C calibration coefficients. This is consistent with
equipment calibration in a mass production environment where
calibration at just a single temperature is practical.
Data Sheet ADL5902
Rev. B | Page 25 of 28
HIGH FREQUENCY PERFORMANCE
The ADL5902 is specified to 6 GHz; however, operation is
possible to as high as 9 GHz with sufficient dynamic range for
many purposes. Figure 52 shows the typical VOUT response and
conformance error at 7 GHz, 8 GHz, and 9 GHz.
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
0
0.25
0.50
0.75
1.00
1.25
1.50
1.75
2.00
2.25
2.50
2.75
3.00
–50 –40 –30 –20 –10 0 10
ERROR (dB)
OUTPUT VOLTAGE (V)
7GHz
8GHz
9GHz
P
IN
(dBm)
08218-057
Figure 52. Typical VOUT and Log Conformance Error at 7 GHz, 8 GHz,
and 9 GHz, 25°C Only
LOW FREQUENCY PERFORMANCE
The lowest frequency of operation of the ADL5902 is approxi-
mately 50 MHz. This is the result of the circuit design and
architecture of the ADL5902.
DESCRIPTION OF CHARACTERIZATION
The general hardware configuration used for most of the ADL5902
characterization is shown in Figure 53. The ADL5902 was driven in
a single-ended configuration for most characterization, except
where noted.
Much of the data was taken using an Agilent E4438C signal source
as a RF input stimulus. Several ADL5902 devices mounted on
circuit boards constructed of Rodgers 3006 material are put into
a test chamber simultaneously, and a Keithley S46 RF switching
network connected the signal source to the appropriate device
under test. The test chamber temperature was set to cycle over
the appropriate temperature range. The signal source, switching,
and chamber temperature are all controlled by a PC running
Agilent VEE Pro.
The subsequent response to stimulus was measured with a
voltmeter and the results stored in a database for analysis later.
In this way, multiple ADL5902 devices are characterized over
amplitude, frequency, and temperature in a minimum amount
of time. The RF stimulus amplitude was calibrated up to the
circuit board that carries the ADL5902, and, thus, it does not
account for the slight losses due to the connector on the circuit
board that carries the ADL5902 nor for the loss of traces on the
circuit board. For this reason, there is a small absolute amplitude
error (generally <0.5 dB) not accounted for in the characterization
data, but this is generally not important because the ADL5902
relative accuracy is unaffected.
ADL5902
CHARACTERIZ ATION
BOARD – TEST SITE 1
ADL5902
CHARACTERIZ ATION
BOARD – TEST SITE 2
KEITHLEY S46
MICROWAVE
SWITCH
AGILENT E8251A
MICROWAVE
SIGNAL
GENERATOR
AGILENT 34980A
SWITCH MATRIX/
DC METER
AGILENT E3631A
DC POWER
SUPPLIES
PERSONA L
COMPUTER
ADL5902
CHARACTERIZ ATION
BOARD – TEST SITE 3
RF DC DATA AND CONTRO L
08218-075
Figure 53. General Characterization Configuration
ADL5902 Data Sheet
Rev. B | Page 26 of 28
EVALUATION BOARD SCHEMATICS AND ARTWORK
X
2
X
2
BIAS AND POWER-
DOWN CONTROL
1
NC
NC
NC
LINEAR-IN-dB VGA
(NEGATIVE SLOPE)
I
DET
26pF
2
3
4
11
10
9
5
6
7
8
16
15
14
13
ADL5902
12
VREF
2.3V
TEMPERATURE
SENSOR
INLO
INHI
VPOS POS
TEMP
(BLACK)
GND
(BLACK)
VSET
(BLACK)
VOUT
(BLACK)
VTGT
(BLACK)
VREF
(BLACK)
TEMP
VSET
VOUT
CLPF
COMM
COMMVTGTVREFTADJ/PWDN
G = 5
I
TGT
R3
60.4
C10
100pF
C12
100pF
V
POS
+5V
(RED)
C4
100pF
C3
0.1µF
C5
100pF
C7
0.1µF
R11
2k
R10
3.74k
R12
301
R9
1430
C9
10µF
RFIN R2
OPEN
R6
0
R1
0
R15
OPEN
TC2 PWDN
(BLACK)
08218-150
Figure 54. Evaluation Board Schematic
Table 8. Evaluation Board Configuration Options
Component Function/Notes Default Value
C10, C12, R3 RF input. The ADL5902 is generally driven single-ended. R3 is the input termination resistor and is
chosen to give a 50 Ω input impedance over a broad frequency range.
C10 = C12 = 100 pF
R3 = 60.4 Ω
R10, R11 VTGT interface. R10 and R11 are set up to provide 0.8 V to VTGT derived from VREF. R10 = 3.74 kΩ,
R11 = 2 kΩ
C4, C5, C7, C3 Power supply decoupling. The nominal supply decoupling consists of two pairs of 100 pF and
0.1 μF capacitors placed close to the two power supply pins of the ADL5902.
C4 = C5 = 100 pF,
C7 = C3 = 0.1 μF
R1, R15, R2,
R6
Output interface. In measurement mode, a portion of the voltage at the VOUT pin is fed back to
the VSET pin via R6. Using the voltage divider created by R2 and R6, the magnitude of the slope
of VOUT is increased by reducing the portion of VOUT that is fed back to VSET. In controller mode, R6
must be open. In this mode, the ADL5902 can control the gain of an external component. A
setpoint voltage is applied to the VSET pin, the value of which corresponds to the desired RF input
signal level applied to the ADL5902.
R1 = R6= 0 Ω,
R2 = R15 = open
C9 Low-pass filter capacitors, CLPF. The low-pass filter capacitor provides the averaging for the
ADL5902 rms computation.
C9 = 0.1 μF
R9, R12 TADJ/PWDN. The TADJ/PWDN pin controls the amount of nonlinear intercept temperature
compensation and/or shuts down the device. The evaluation board is configured with TADJ
connected to VREF through a resistor divider (R9, R12).
R9 = 1430 Ω
R12 = 301 Ω
. o a 'VPOS' 06m). 0 o . . . . . . . . . 2 41101.. ._vm, .IEMP, ,vsn. .. ' n10. . - . . gNéLop ' DEVICES . MDLSQOZ'EVALIZ-RUI B u 0'.
Data Sheet ADL5902
Rev. B | Page 27 of 28
ASSEMBLY DRAWINGS
08218-060
Figure 55. Evaluation Board Layout, Top Side
08218-061
Figure 56. Evaluation Board Layout, Bottom Side
Sg‘filéécsi www.3nalng.cnm
ADL5902 Data Sheet
Rev. B | Page 28 of 28
OUTLINE DIMENSIONS
*COMPLIANT
TO
JEDEC STANDARDS MO-220-WGGC-3
WITH EXCEPTION TO THE EXPOSED PAD.
1
0.65
BSC
16
5
8
9
12
13
4
PIN 1
INDICATOR
4.10
4.00 SQ
3.90
0.50
0.40
0.30
SEATING
PLANE
0.80
0.75
0.70 0.05 MAX
0.02 NOM
0.20 REF
0.25 MIN
COPLANARITY
0.08
PIN 1
INDICATOR
0.35
0.30
0.25
*2.40
2.35 SQ
2.30
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
07-21-2015-B
BOTTOM VIEW
TOP VIEW
EXPOSED
PAD
PKG-000000
Figure 57. 16-Lead Lead Frame Chip Scale Package [LFCSP]
4 mm × 4 mm Body and 0.75 mm Package Height
(CP-16-20)
Dimensions shown in millimeters
ORDERING GUIDE
Model1 Temperature Range Package Description Package Option Ordering Quantity
ADL5902ACPZ-R7 −40°C to +125°C 16-Lead Lead Frame Chip Scale Package [LFCSP] CP-16-20 1,500
ADL5902ACPZ-R2 −40°C to +125°C 16-Lead Lead Frame Chip Scale Package [LFCSP] CP-16-20 250
ADL5902ACPZ-WP −40°C to +125°C 16-Lead Lead Frame Chip Scale Package [LFCSP] CP-16-20 64
ADL5902-EVALZ Evaluation Board
1 Z = RoHS Compliant Part.
©2010–2016 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D08218-0-8/16(B)

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