INA28x User Guide Datasheet by Texas Instruments

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i EXAS INSTRUMEN «m Tl EZE'“ Cummunity
Protection
VOUT
V+
5V DC
+
Load (for example, Motor)
RSH
VBUS
–14 V to 80 V
INA282-286
Motor
TI Designs
EMC Compliant High Side Current Sensing with
Overvoltage Protection
TI Designs Design Features
TI Designs provide the foundation that you need Wide common mode input range: –14 V to 80 V
including methodology, testing and design files to Overvoltage protection: 45 V
quickly evaluate and customize and system. TI EFT protection up to 1 kV as per IEC61000-4-4
Designs help you accelerate your time to market. > 130 dB Common Mode Rejection Ratio (CMRR)
Design Resources for DC-10 Hz
Overall accuracy better than 2%
TIDA-00126 Design Files 70-µV offset and 1.4% gain error
INA282 Product Folder
Featured Applications
Factory automation: PLC 24-V DC bus current
ASK Our Analog Experts monitoring
WebBench Calculator Tools 24-V system/board level current sensing
Bi-directional motor control
Smart battery packs and chargers
Solar inverters
28-V auxiliary input current measurement for
aerospace
Electric hybrid vehicles
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An IMPORTANT NOTICE at the end of this TI reference design addresses authorized use, intellectual property matters and other
important disclaimers and information.
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Overview
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1 Overview
High-side precision current sensing is widespread - from industrial equipment like protection relays,
solenoid or motor control, test equipment and solar inverters to consumer equipment like smart phones,
tablets, servers and battery chargers. Engines can use the amount of current being delivered to a load to
make safety-critical decisions and avoid failures due to overcurrent or short-circuit conditions by
maintaining the load current within safe operating limits.
This reference design focuses on EMC-compliant high-side current sense solutions using the INA282,
INA283, INA284, INA285, and INA286 family of voltage output current shunt monitor devices. These
devices help designers achieve highly-accurate current-monitoring solutions in a wide range of common-
mode voltages from –14V to +80V. This device family also supports bi-directional operation that may be
required in battery operated equipment where charging and discharging currents need to be monitored.
Clearly, these devices are likely to encounter very high and dynamic changes in common mode voltages
when accessing their power supplies. This ability is useful in applications when current shunt monitor
devices must interface with a low-voltage analog-to-digital converter (ADC). In such a scenario, both the
current shunt monitor device and the ADC can be powered with the same supply voltage regardless of the
system’s common-mode voltage.
2 Design Specifications
The high-side current sense is designed to meet the following specifications:
Load supply up to 24 V
Overvoltage protection up to 45 V
Device supply voltage of 5 VDC
1 kV electrical fast transient (EFT) withstanding capability
Overall accuracy better than 2%
3 Circuit Diagram
A circuit diagram of high-side current sensing with improved transient immunity is shown in Figure 1.
Figure 1. High-Side Current Sensing with Improved Transient Immunity
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Device Gain (V/V)
INA282 50
INA286 100
INA283 200
INA284 500
INA285 1000
System Sourcing
System Sinking
Differential
Amplifier
Vref
Vref
Vo
Vo
Rshunt
Sshunt
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GND
Vbus
GND
+
+
System
Load
+
www.ti.com
Theory of Operation
4 Theory of Operation
The system implementing high-side current sensing puts a shunt resistor between the supply voltage
(VBUS) and the load. High-side current sensing is desirable as any downstream failure can be detected and
appropriate corrective action can be triggered. High-side current sensing can be seen as a small sense
voltage riding on top of a high common mode voltage. That is why high-side current shunt monitors must
have a common mode voltage range outside the load’s supply voltage and a very high CMRR. Current
sense monitors encounter high voltage transients and overvoltage events frequently in the fields.
Transient voltage can cause severe damage and failure of the device. Overcoming unwanted damaging
transient threats is one of the biggest challenges in the design. Therefore, adding robust EMC protection
externally becomes a necessity. The EMC protection circuit should protect the device from the transient
high voltages and maintain stable output to keep the circuit working even when transient conditions occur.
The INA282-286 devices are voltage output, high-side measurement, unidirectional and bi-directional, and
zero-drift current shunt monitors. This family of devices has predetermined gains that range from 50 V/V to
1000 V/V. The corresponding gain of the specific device amplifies the voltage developed across the
device inputs. The output pin presents the voltage. The INA282-286 devices can sense voltage drops
across shunts at common-mode voltages between –14 V to 80 V, independent of supply voltages and 140
dB CMRR (Typical). These devices operate with supply voltages between 2.7 V and 18 V and draw a
maximum of 900-μA supply current. The INA282-286 devices are used for accurate measurements well
outside of their own power-supply voltages (V+). For example, the V+ power supply can be 5 V while the
common-mode voltage may be as high as +80 V.
The output of the device is proportional to the current through the sense resistor:
VOUT = (GAIN × RSH × ILOAD)+VREF = GAIN × VSH + VREF
Where, VREF is the average of VREF1 and VREF2.
Note: VREF1 and VREF2 control the VOUT level for bi-directional operation. Make sure VREF is sufficiently high
such that output voltage does not exceed the allowed output swing of the device. The output voltage
swings above VREF for positive sense current direction and below VREF for negative sense current direction.
The output voltage stays at VREF when VSH is zero.
For unidirectional current sensing, REF1 and REF2 pins connect to the ground. Then, represent output
voltage as:
VOUT = GAIN × RSH × ILOAD = GAIN × VSH
Figure 2. Typical High Side Current Sensing
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l TEXAS INSTRUMENTS SH
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4.1 Sizing Shunt Resistor (RSH)
Selection of the correct sense resistor is vital for accurate current measurement in an application. To
determine the size of the shunt resistors, the following parameters must be known:
Full scale load current
Available supply voltage for the device (V+ = 5 VDC)
Minimum load voltage requirement (or maximum permissible voltage loss in the measurement line)
• Accuracy
Resolve trade-offs while selecting and calculating the optimum value of RSH:
Increasing RSH increases the VSH, which versus A large RSH value increases the I2× R losses
provides better accuracy because voltage which in-turn increases self-heating and
offset and input bias current errors become changes the value of RSH and also causes
less significant. higher voltage loss that must meet the load’s
minimum voltage requirement.
Increasing RSH increases the VSH which must versus The minimum value of RSH is set by input
not exceed the input voltage swing specified dynamic range, input offset voltage, and
by the device. resolution requirements.
Tighter tolerance, low TCR, low thermal versus Cost
EMF, 2-pin or 4-pin sense resistor, all need a
very low inductance resistor if the current
being sensed contains high-frequencies.
(Wire-wound resistors have higher
inductance compared to metal-film resistors.)
Step 1: Output Voltage Swing
Find the output voltage swing from the device datasheet, which is:
(GND + 0.4 V) < VOUT < (V+ – 0.4 V); where V+ is 5 VDC
0.4 V < VOUT < 5 V – 0.4 V
Output voltage swing: 0.4 V < VOUT < 4.6 V
Step 2: Input Sense Voltage Range
Refer the above relation to input by dividing it with device gain. For the INA282 device, the gain is 50
V/V.
Input sense voltage (VSH) range: 800 µV < VSH < 92 mV for the given power supply (V+) of 5 V
Step 3: Maximum Sense Resistor
If a peak load current of 0.8 A is expected in an application and the maximum input sense voltage
VSH (MAX) must not exceed 92 mV, use this formula:
(1)
Choose a value for the reference design: RSH (MAX) = 100 mΩ.
Note: For most applications, the best performance is attained with an RSH value that provides a full-
scale sense voltage.
Step 4: Minimum Load Current
Find the minimum load current IL (MIN):
Either the total error budget of the device or the minimum input sense voltage VSH (MIN) = 800 µV
(whichever is more) limits the minimum load current (IL (MIN)) that can be accurately represented by the
INA282.
(2)
IL (MIN) = 8 mA
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Rpp Rpn
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High Current Path High Current Path High Current Path
www.ti.com
Theory of Operation
So, the minimum load current (IL (MIN)) producing change in the output voltage is greater than or equal to
8 mA.
Step 5: Maximum Power Dissipation
Maximum Power Dissipation: PSH (MAX) = IL (MAX)2× RSH = (0.8 A × 0.8 A) × 100 mΩ
PSH (MAX) = 64 mW
Select a sense resistor having maximum power dissipation more than 64 mW.
Note: If the engineer allows the sense resistor to dissipate more power, the sense resistor heats up
and its maximum power distribution value drifts.
Step 6: Voltage Loss
Find the maximum voltage loss caused by the sense resistor using(RSH) using this formula:
Maximum voltage loss = VSH (MAX) = 80 mV
Example: If the VBUS = 24 V, then the minimum voltage delivered to the load is:
VL (MIN) = VBUS – VSH (MAX) = 24 V – 0.080 V = 23.92 V
Make sure the minimum voltage delivered meets the minimum voltage requirement of the load.
4.2 Recommended PCB Layout for RSH
Be aware of PCB layout parasitic:
Always ensure that the sense resistor is Kelvin-connected.
Make the input traces as short as possible.
Make the input traces as balanced as possible.
Place the current sensing device and shunt on the same side of the PCB.
To determine an error contributed by device, measure the voltage across device pins not across the
sense resistor.
4.3 Transient Protection
In industrial and automotive environments, electronic devices can be subjected to wide input voltage
variations resulting from operating relays, solenoid switching, inductive load kick-back, load dump pulses,
and reverse polarity. A load dump condition occurs when the load from the generator delivering current is
abruptly disconnected. A load dump condition can be up to +80 V. Battery polarity reversal causes a
negative input of common mode voltage up to –12 V. In the event the device is exposed to transients on
input in excess of its ratings, then external transient absorbers (zener or TVS diodes) are required. The
TVS safeguards sensitive devices and common circuitry by clamping the voltage level and diverting
transient currents when a trigger voltage is reached. This design uses two unidirectional transient voltage
suppressors in series with opposite polarities on VIN+ and VIN– pins to take care of the asymmetrical
common mode voltage rating of the device. The two series opposite zener diode D1 and D2 placed
between the differential inputs of INA282 make sure the differential input voltage never exceeds its
absolute maximum rating of ±5 V.
(3)
When VCis the clamping voltage of TVS at IP:
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PC(MAX) BR(MAX) BR(MAX)
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ZSis the source impedance of the EFT pulse generator and RSis the external series filter resistance.
ZS= 50 Ωand RS= 10 Ω.
(4)
Pulse Power Dissipation = PP= VC× IP(5)
D3 and D5 TVS Selection:
Usually select a TVS diode having a stand-off voltage or working voltage greater than the maximum
expected VBUS so that the TVS does operate or interfere during the normal operation. For PLC
applications, the 24-V supply may go up to 20% higher than 24-V nominal supply voltage. Any positive
transient voltages are quickly clamped below 80 V.
This is why 28.8 V < VRand VC (MAX) < 80 V (Maximum common mode voltage rating of the device).
D4 and D6 TVS Selection:
For any negative transient voltages, select a TVS diode that clamps before reaching 14 V.
Once the primary selection is done, solve equations 1, 2, and 3 to find-out the following parameters which
are important for TVS selection:
Pulse current (IP) flowing through the TVS
Clamping voltage (VC) across the TVS at IP
Pulse power (PPP) in the TVS at VCand IP
Make sure clamping voltage across D3 and D5 does not exceed 80 V (in fact, the clamping voltage should
be well within 80 V) during a 1-kV positive fast transient event. Likewise, make sure clamping the voltage
across D4 and D6 does not exceed 14 V (in fact, should be well within 14 V) during 1-kV negative fast
transient event.
Make sure the pulse power dissipation in any of the TVSs exceed their maximum allowed peak pulse
power dissipation ratings.
To perform the transient protection job, the following TVS diodes have been selected:
D3 and D5:
SMBJ40A (Rating: VR= 40 V, VC (MAX) = 64.5 V at IPP (MAX) = 9.3 V and PPP (MAX) = 600 W at 10/1000 µs or
greater than 10 kW at 5/50 ns).
Use the TVS ratings; solve for equations 1, 2, and 3 for IP, VCand PP:
IP= 15.42 A
VC= 74.7 V, which is less than 80 V common mode voltage rating of INA282.
PP= VCX IP= 1152 W
D4 and D6: SMAJ7.0A (Rating: VR=7V,VC (MAX) =12VatIPP (MAX) = 33.3 V and PPP (MAX) = 400 W at
10/1000 µs or greater than 10 kW at 5/50 ns)
Use the TVS ratings; solve for equations 1, 2, and 3 for IP, VCand PP:
IP= 16.5 A
VC= 10.3 V, which is less than 14-V common mode voltage rating of INA282.
PP= VC× IP= 170 W
The rise and fall time for an EFT pulse are 5 ns and 50 ns, respectively, as illustrated in Figure 3. The
pulse width is 55 ns (less than 0.1 µs).
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l TEXAS INSTRUMENTS RI % 7 7 IN+ F‘LTER + FILTER
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www.ti.com
Theory of Operation
Figure 3. EFT Pulse
The SMBJ40A and SMAJ7.0A transient voltage suppressors in the design have peak pulse power ratings
of 600 W and 500 W, respectively, when tested with a convention of 10/1000 µs double exponential
waveform. The TVS manufacturer provides a peak pulse power versus pulse time graph, which shows
how a shorter or longer duration affects the peak pulse power of a TVS. For shorter pulse widths, TVS can
withstand higher peak pulse power. Therefore, for 5/50 ns EFT pulses, SMBJ40A and SMAJ7.0A transient
voltage suppressors can sustain more than 10 kW peak pulse power.
4.4 Input Filter
TI placed an EMI/RFI filter network between the sense resistor and the INA282 device input pins to reject
any ac noise, fast transients and current spikes. EFT bursts is a wideband phenomenon with spectral
components up to hundreds of MHz. EFT bursts appear as common mode pulses to the high side current
shunt monitor devices. The input filter uses RC components to provide both common-mode and
differential filtering. The common mode filter uses 0.033 µF/2 kV Y-Cap to take care of high voltage high
frequency common mode transients (EFT bursts).
The differential filter cut-off frequency is calculated as:
(6)
FDMC = 9.6 kHz (approximately)
Adding any external filter resistor in series with the current shunt monitor’s input will cause additional gain
error and degrade CMR due to resistance value mismatch.
(7)
RIN is the internal input impedance of the INA282 current shunt monitor.
If the inputs use a pair of 10-Ω, 1% resistors, additional gain error will be 0.1664%. To ensure better
accuracy, the filter resistor should be less than or equal to 10 Ω. The engineer can also determine the
worst-case gain error by inserting extreme tolerances of RFILTER and RIN in the above equation. Therefore,
the filter resistor must have 1% tolerance or better.
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l TEXAS INSTRUMENTS mar am "/- (SBOS485) total Error vs. nmerenual Innuannaln um w. m , , m , Mum-Emu _ um um _wuvrm m _rmv;m _rmv.s% nwmaonusawruwwrw nmemu-Ivnlugumv)
Theory of Operation
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4.5 Source of Errors
The following list includes all the possible error sources:
Input offset voltage
Input offset voltage drift with temperature
Input offset voltage drift with time
Input offset current
Gain error
Linearity error
Common mode rejection
Power supply rejection
Sense resistor tolerance
Reference common mode rejection
Addition gain error due to external filter resistance mismatch
Refer to the CALCULATING TOTAL ERROR section of the INA282 datasheet (SBOS485) for information
about how these errors affect the overall accuracy.
For small differential signals at the input, the error
is dominated by the amplifier’s offset voltage. Low
input offset is critical to achieving accurate
measurements at the low end of the dynamic
range.
For large differential signals at the input, the error is
dominated by the amplifier’s gain error.
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l TEXAS INSTRUMENTS EUT (INAzszzasEx/M wnh added aanery m eeneme Decade Luau Genevamr
EUT (INA282-286EVM with added
EMC Protection)
Battery to Generate
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EFT
Capacitive
Clamp
EFT Burst
Generator
DMM-2 for
VSENSE
DMM-1 for
VOUT
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EFT Test Setup
5 EFT Test Setup
Example EFT test setups are illustrated in Figure 4 and Figure 5.
Figure 4. EFT Test Setup View 1
Figure 5. EFT Test Setup View 2
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( )
V  GAIN V
Measured Output Voltage Theoretical Output Voltage OUT SH  1 00
Theoretical
% VOU  Output Voltage GAIN
T VSH
-
´
D
´
´= -=
Pre-Compliance EFT Test Results
www.ti.com
The EFT test setup consists of:
Two 6½ digital multi-meters (DMMs) One DMM measures VSH and other measures VOUT
EUT Modified INA282-286EVM
Battery Provides 5 VDC supply to INA282 device
24-V regulated power supply Used as 24-V load supply
EFT burst generator Generates the 1 kV, 5 kHz and 100 kHz EFT burst pulses for 1 minute duration
Capacitive clamp To couple EFT pulses to the EUT as a common mode input voltage
Decade box load Used to set the desired load value
The two DMMs are put in MIN-MAX mode to capture and log the minimum and maximum excursions on
VSH and VOUT during the application of EFT pluses. After the test is complete, minimum and maximum
values of VSH and VOUT are retrieved from the DMMs. Later these values are used to calculate the
accuracy.
6 Pre-Compliance EFT Test Results
The design has been implemented utilizing an existing high-side current monitor evaluation module
(INA282-286EVM) and the module was modified to add the low pass filter, zener diodes and TVS to meet
the EFT bursts test as per IEC61000-4-4. The output voltage accuracy can be calculated as:
(8)
Test Conditions: the following conditions apply to the results shown in Table 1 and Table 2: VBUS = 24
VDC, V+ = 5 VDC, Device used is INA282, GAININA282 = 50 V/V, Load resistance = 30.0 Ωand RSH = 0.1 Ω
at 25°C.
Table 1. Test 1
Test Name/Condition Shunt Voltage (VSH) Output Voltage (VOUT) % Error
Functional 80.870 mV 4.0390 V 0.1113%
Table 2. Test 2 EFT Burst
Test Name/Condition Shunt Voltage (VSH) Output Voltage (VOUT) % Error
1000 V, 100 kHz, negative 80.471441 mV 4.069547 V 1.143%
pulses
1000 V, 100 kHz, positive 81.10343 mV 4.017819 V 0.9211%
pulses
1000 V, 5 kHz, negative pulses 80.93304 mV 4.036456 V 0.252%
1000 V, 5 kHz, positive pulses 80.99869 mV 4.036590 V 0.33%
7 Conclusion
The reference design presents the details for designing high-side current shunt monitors with EMC
protection that meets an overall accuracy of 2%. Adding an external filter to the current shunt monitor
might degrade the performance unless designed with appropriate considerations. TI offers INA282
voltage-output with high-side current sense monitors. These monitors solve the common and often
challenging problem of measuring high-side current, especially when common-mode dynamics go
negative below ground. The common-mode voltage range for the INA282 is independent of the supply
voltage. The zero-drift architecture, unique input stage topology, and the precisely trimmed internal
resistor of the INA282 experiences very low offset voltage and offset drift over temperature and time that
is crucial to maintain accuracy in high voltage applications with a high degree of dynamic changes in
common-mode voltage.
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IC CURRENT SHUNT MONITOR 8SOIC
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8SOIC
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP
IC CURRENT SHUNT MONITOR 8VSSOP