ST1S41 Datasheet by STMicroelectronics

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This is information on a product in full production.
April 2013 DocID023654 Rev 2 1/27
27
ST1S41
4 A step-down switching regulator
Datasheet - production data
Features
4 A output current
4.0 V to 18 V input voltage
Output voltage adjustable from 0.8 V
850 kHz switching frequency
Internal soft-start
Integrated 95 mΩ and 69 mΩ power
MOSFETs
All ceramic capacitor
Enable
Cycle-by-cycle current limiting
Current foldback short-circuit protection
VFQFPN 4x4-8L and HSOP-8 packages
Applications
•μP/ASIC/DSP/FPGA core and I/O supplies
Point of load for: STB, TVs, DVD
Optical storage, hard disk drive, printers,
audio/graphic cards
Description
The ST1S41 is an internally compensated 850
kHz fixed-frequency PWM synchronous step-
down regulator. The ST1S41 operates from 4.0 V
to 18 V input, while it regulates an output voltage
as low as 0.8 V and up to VIN.
The ST1S41 integrates 95 mΩ high-side switch
and 69 mΩ synchronous rectifier allowing very
high efficiency with very low output voltages.
The peak current mode control with internal
compensation delivers a very compact solution
with a minimum component count.
The ST1S41 is available in VFQFPN 4 mm x 4
mm 8-lead package and HSOP-8.
Figure 1. Application circuit
VFQFPN8 4x4 HSOP8
VINSW SW
FB
PGND
EN
ST1S41
Cin_sw
Cout
L
R1
R2
VINA
Cin_a
VOUT
VIN
ePAD/GND
1
48
2
3
67
9
AM15058v1
www.st.com
Contents ST1S41
2/27 DocID023654 Rev 2
Contents
1 Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.1 Pin connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.2 Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
2 Maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
3 Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
4 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
5 Functional description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
5.1 Internal soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
5.2 Error amplifier and control loop stability . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
5.3 Overcurrent protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
5.4 Enable function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
5.5 Hysteretic thermal shutdown . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
6 Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
6.1 Input capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
6.2 Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
6.3 Output capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
6.4 Thermal dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
6.5 Layout considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
7 Typical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
8 Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
9 Ordering information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
10 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
DocID023654 Rev 2 3/27
ST1S41 Pin settings
1 Pin settings
1.1 Pin connection
Figure 2. Pin connection (top view)
1.2 Pin description
1
45
8
VINA
EN
FB
GND
PGND
SW
VINSW
NC
EN
NC
VFQFPN HSOP8
2
3
6
7
9
9
AM15059v1
Table 1. Pin description
N. Type Description
1 VINA Unregulated DC input voltage
2EN
Enable input. With EN higher than 1.2 V the device in ON and with EN
lower than 0.4 V the device is OFF (ST1S41Ixx).
3FB
Feedback input. Connecting the output voltage directly to this pin the
output voltage is regulated at 0.8 V. To have higher regulated voltages
an external resistor divider is required from Vout to FB pin.
4 AGND Ground
5 NC Can be connected to ground
6 VINSW Power input voltage
7 SW Regulator output switching pin
8 PGND Power ground
9 ePAD Ground
Maximum ratings ST1S41
4/27 DocID023654 Rev 2
2 Maximum ratings
3 Thermal data
Table 2. Absolute maximum ratings
Symbol Parameter Value Unit
VINSW Power input voltage -0.3 to 20
V
VINA Input voltage -0.3 to 20
VEN Enable voltage -0.3 to VINA
VSW Output switching voltage -1 to VIN
VPG Power Good -0.3 to VIN
VFB Feedback voltage -0.3 to 2.5
IFB FB current -1 to +1 mA
PTOT Power dissipation at TA < 60 °C 2.25 W
TOP Operating junction temperature range -40 to 150 °C
Tstg Storage temperature range -55 to 150 °C
Table 3. Thermal data
Symbol Parameter Value Unit
RthJA Maximum thermal resistance
junction-ambient (1)
1. Package mounted on demonstration board.
VFQFPN 40 °C/W
HSOP8 40
é]
DocID023654 Rev 2 5/27
ST1S41 Electrical characteristics
4 Electrical characteristics
TJ = 25 °C, VCC = 12 V, unless otherwise specified.
Table 4. Electrical characteristics
Symbol Parameter Test condition Values Unit
Min. Typ. Max.
VIN Operating input voltage
range (1) 418
V
VINON Turn-on VCC threshold (1) 2.9
VINHYS Threshold hysteresis (1) 0.250
RDSON-P High-side switch on-
resistance ISW=750 mA 95 mΩ
RDSON-N Low-side switch on-
resistance ISW=750 mA 69 mΩ
ILIM Maximum limiting current (2) 5.0 7.0 A
Oscillator
FSW Switching frequency 0.7 0.85 1 MHz
DMAX Maximum duty cycle (2) 100 %
Dynamic characteristics
VFB Feedback voltage 0.784 0.8 0.816 V
(1) 0.776 0.8 0.824
%VOUT/
ΔIOUT
Reference load
regulation Isw=10 mA to ILIM (2) 0.5 %
%VOUT/
ΔVIN Reference line regulation VIN= 4.0 V to 18 V (2) 0.4 %
DC characteristics
IQQuiescent current Duty cycle=0, no load
VFB=1.2 V 1.5 2.5 mA
IQST-BY Total standby quiescent
current
OFF 2.4 4.5 μA
OFF (1) 6
IFB FB bias current 50 nA
Enable
VEN EN threshold voltage Device ON level 1.2 V
Device OFF level 0.4
IEN EN current 2 μA
Soft-start
TSS Soft-start duration 1 ms
Electrical characteristics ST1S41
6/27 DocID023654 Rev 2
Protection
TSHDN
Thermal shutdown 150 °C
Hystereris 15
1. Specifications referred to TJ from -40 to +125 °C. Specifications in the -40 to +125 °C temperature range
are assured by design, characterization and statistical correlation.
2. Guaranteed by design.
Table 4. Electrical characteristics (continued)
Symbol Parameter Test condition Values Unit
Min. Typ. Max.
é]
DocID023654 Rev 2 7/27
ST1S41 Functional description
5 Functional description
The ST1S41 is based on a “peak current mode”, constant frequency control. The output
voltage VOUT is sensed by the feedback pin (FB) compared to an internal reference (0.8 V)
providing an error signal that, compared to the output of the current sense amplifier, controls
the on and off-time of the power switch.
The main internal blocks are shown in the block diagram in Figure 3. They are:
A fully integrated oscillator that provides the internal clock and the ramp for the slope
compensation avoiding sub-harmonic instability
The soft-start circuitry to limit inrush current during the startup phase
The transconductance error amplifier with integrated compensation network
The pulse width modulator and the relative logic circuitry necessary to drive the internal
power switches
The drivers for embedded P-channel and N-channel power MOSFET switches
The high-side current sensing block
The low-side current sense to implement diode emulation
A voltage monitor circuitry (UVLO) that checks the input and internal voltages
A thermal shutdown block, to prevent thermal run-away.
Figure 3. Block diagram
OSC
E/A
DRIVER
DRIVER
DMD
OTP
MOSFET
CONTROL
LOGIC
REGULATOR
SHUTDOWN
I_SENSE
COMP
COMP
OCP
REF
0.8V
SOFT-START
Vsum
Vc
OCP
UVLO
Vdrv_p
Vdrv_n
I2V RSENSE
VINA VINSW
SW
GNDPGNDAENFB
AM15060v1
Functional description ST1S41
8/27 DocID023654 Rev 2
5.1 Internal soft-start
The soft-start is essential to assure correct and safe startup of the step-down converter. It
avoids inrush current surge and makes the output voltage increase monothonically.
The soft-start is performed by ramping the non-inverting input (VREF) of the error amplifier
from 0 V to 0.8 V in around 1 ms.
5.2 Error amplifier and control loop stability
The error amplifier compares the FB pin voltage with the internal 0.8 V reference and it
provides the error signal to be compared with the output of the current sense circuitry, that is
the high-side power MOSFET current. Comparing the output of the error amplifier and the
peak inductor current implements the peak current mode control loop.
The error amplifier is a transconductance amplifier (OTA). The uncompensated
characteristics are listed in Table 5:
The ST1S41 embeds the compensation network that assures the stability of the loop in the
whole operating range. Here below, all the tools needed to check the loop stability.
In Figure 4 the simple small signal model for the peak current mode control loop is shown.
Table 5. Error amplifier characteristics
DC gain 95 dB
Gm 251 uA/V
Ro 240 MΩ
v‘
DocID023654 Rev 2 9/27
ST1S41 Functional description
Figure 4. Block diagram of the loop for the small signal analysis
Three main terms can be identified to obtain the loop transfer function:
1. from control (output of E/A) to output, GCO(s);
2. from output (Vout) to FB pin, GDIV(s);
3. from FB pin to control (output of E/A), GEA(s).
The transfer function from control to output GCO(s) results:
Equation 1
where RLOAD represents the load resistance, Ri the equivalent sensing resistor of the
current sense circuitry, ω p the single pole introduced by the LC filter and ω z the zero given
by the ESR of the output capacitor.
FH(s) accounts for the sampling effect performed by the PWM comparator on the output of
the error amplifier that introduces a double pole at one half of the switching frequency.
Equation 2
L
Cout
Current sense
Logic
and
Driver
Slope
Compensation
PW M com parator
Error Amp
Rc
Cc
R1
R2
0.8V
High side
Switch
Low side
Switch
GCO(s)
GDIV(s )
GEA(s)
VIN
VC
VOUT
VFB
AM15061v1
GCO s() RLOAD
Ri
------------------1
1Rout TSW
L
------------------------------ mC1D()0.5[]+
-----------------------------------------------------------------------------------------------------
1s
ω
z
-------+
⎝⎠
⎛⎞
1s
ω
p
-------+
⎝⎠
⎛⎞
---------------------- FHs()⋅⋅=
ω
Z 1
ESR COUT
----------------------------------=
Functional description ST1S41
10/27 DocID023654 Rev 2
Equation 3
where:
Equation 4
Sn represents the ON time slope of the sensed inductor current, Se the slope of the external
ramp (VPP peak-to-peak amplitude) that implements the slope compensation to avoid sub-
harmonic oscillations at duty cycle over 50%.
The sampling effect contribution FH(s) is:
Equation 5
where:
Equation 6
and
Equation 7
The resistor to adjust the output voltage gives the term from output voltage to the FB pin.
GDIV(s) is:
Equation 8
ω
p 1
RLOAD COUT
---------------------------------------- mC1D()0.5
LC
OUT fSW
⋅⋅
------------------------------------------------+=
mC1Se
Sn
------ +=
SeVpp fSW
=
Sn
VIN VOUT
L
------------------------------Ri
=
FHs() 1
1s
ω
n QP
---------------------- s2
ω
n
2
-------++
----------------------------------------------=
QP1
πmC1D()0.5[]
----------------------------------------------------------------=
ω
n πfSW
=
GDIV s() R2
R1R2
+
--------------------=
E] 44PM
DocID023654 Rev 2 11/27
ST1S41 Functional description
The transfer function from FB to Vc (output of E/A) introduces the singularities (poles and
zeroes) to stabilize the loop. Figure 5 shows the small signal model of the error amplifier
with the internal compensation network.
Figure 5. Small signal model for the error amplifier
RC and CC introduce a pole and a zero in the open loop gain. CP does not significantly affect
system stability and can be neglected.
So GEA(s) results:
Equation 9
where GEA= Gm · Ro.
The poles of this transfer function are (if Cc >> C0+CP):
Equation 10
Equation 11
whereas the zero is defined as:
Equation 12
The embedded compensation network is RC=70 kΩ, CC=195 pF while CP and CO can be
considered as negligible. The error amplifier output resistance is 240 MΩ so the relevant
singularities are:
Equation 13
CoRo
Cc
Rc Cp
Gm
*
Vd
VFB
VREF
Vd
AM15062v1
GEA s() GEA0 1s+RcCc
⋅⋅()
s2R0C0Cp
+()RcCcsR
0Cc
R0C0Cp
+()RcCc
++()1++⋅⋅ ⋅
---------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------=
fP LF 1
2πR0Cc
⋅⋅ ⋅
------------------------------------------=
fP HF 1
2πRcC0Cp
+()⋅⋅ ⋅
------------------------------------------------------------=
fZ1
2πRcCc
⋅⋅ ⋅
------------------------------------------=
fZ11.6 kHz fPLF 3.4 Hz==
Functional description ST1S41
12/27 DocID023654 Rev 2
So closing the loop, the loop gain GLOOP(s) is:
Equation 14
Example:
VIN=12 V, VOUT=1.2 V, Iomax=4 A, L=1.5 uH, Cout=47 uF (MLCC), R1=10 k
Ω
, R2=20
k
Ω
(see Section 6.2 and Section 6.3 for inductor and output capacitor selection
guidelines).
The module and phase bode plot are reported in Figure 6.
The bandwidth is 100 kHz and the phase margin is 45 degrees.
GLOOP s() GCO s() GDIV s() GEA s()⋅⋅=
PM: 1n 1m 1 1 anmy [Hz] 1 10‘ 1mj 1 10‘ 1,5 >41 -72: 11” ~15: 71:25 -2111 u 1 1 In 100 1 1 PM“! 111x] mum
DocID023654 Rev 2 13/27
ST1S41 Functional description
Figure 6. Module and phase bode plot
5.3 Overcurrent protection
The ST1S41 implements the pulse-by-pulse overcurrent protection. The peak current is
sensed through the high-side power MOSFET and when it exceeds the first overcurrent
threshold (OCP1) the high-side is immediately turned off and the low-side conducts the
inductor current for the rest of the clock period and the following high-side cycle is disabled.
This implements a division by two of the switching frequency in case of overload to keep the
output current limited below the current limit value.
During overload condition, since the duty cycle is not set by the control loop but is limited by
the overcurrent threshold, the output voltage drops out of regulation. If the feedback falls
below 0.3 V, the switching frequency is reduced to one fourth and the current limit threshold
is folded back to around 2 A. Thanks to the current and frequency foldback the stress on the
device and on the external power components is reduced in case of severe overload or
dead-short to ground of the output.
Functional description ST1S41
14/27 DocID023654 Rev 2
The current foldback is disabled during the startup to allow the Vout to start up properly in
case of a big output capacitor requiring high extra current to be charged.
A further mechanism is protecting the device in case of short-circuit on the output and high
input voltage. A further threshold (OCP2, 1 A higher than OCP1) is compared to the inductor
current. If the inductor current exceeds OCP2, the device stops switching and it restarts with
a soft-start cycle.
5.4 Enable function
The enable feature allows the device to be put into standby mode. With the EN pin lower
than 0.4 V, the device is disabled and the power consumption is reduced to less than 15 uA.
With the EN pin higher than 1.2 V, the device is enabled. If the EN pin is left floating, an
internal pull-down ensures that the voltage at the pin reaches the inhibit threshold and the
device is disabled. The pin is also VIN compatible.
5.5 Hysteretic thermal shutdown
The thermal shutdown block generates a signal that turns off the power stage if the junction
temperature goes above 150 °C. Once the junction temperature goes back to about 130 °C,
the device restarts in normal operation.
é]
DocID023654 Rev 2 15/27
ST1S41 Application information
6 Application information
6.1 Input capacitor selection
The capacitor connected to the input must be capable of supporting the maximum input
operating voltage and the maximum RMS input current required by the device. The input
capacitor is subject to a pulsed current, the RMS value of which is dissipated over its ESR,
affecting the overall system efficiency.
So the input capacitor must have an RMS current rating higher than the maximum RMS
input current and an ESR value compliant with the expected efficiency.
The maximum RMS input current flowing through the capacitor can be calculated as:
Equation 15
where Io is the maximum DC output current, D is the duty cycle, η is the efficiency.
Considering η =1, this function has a maximum at D=0.5 and is equal to Io/2.
The peak-to-peak voltage across the input capacitor can be calculated as:
Equation 16
where ESR is the equivalent series resistance of the capacitor.
Given the physical dimension, ceramic capacitors can meet well the requirements of the
input filter sustaining a higher input RMS current than electrolytic / tantalum types. In this
case the equation of CIN as a function of the target peak-to-peak voltage ripple (VPP) can be
written as follows:
Equation 17
neglecting the small ESR of ceramic capacitors.
Considering η =1, this function has its maximum in D=0.5, therefore, given the maximum
peak-to-peak input voltage (VPP_MAX), the minimum input capacitor (CIN_MIN) value is:
Equation 18
IRMS IOD2D
2
η
-----------------D2
η
2
-------+=
VPP
IO
CIN FSW
---------------------------- 1D
η
----
⎝⎠
⎛⎞
DD
η
----1D()+ESR IO
+=
CIN
IO
VPP FSW
----------------------------- 1D
η
----
⎝⎠
⎛⎞
DD
η
----1D()+=
CIN_MIN
IO
2V
PP_MAX FSW
⋅⋅
-----------------------------------------------------=
Application information ST1S41
16/27 DocID023654 Rev 2
Typically, CIN is dimensioned to keep the maximum peak-to-peak voltage ripple in the order
of 1% of VINMAX.
In Table 6 some multi-layer ceramic capacitors suitable for this device are reported.
A ceramic bypass capacitor, as close as possible to the VINA pin, so that additional parasitic
ESR and ESL are minimized, is suggested in order to prevent instability on the output
voltage due to noise. The value of the bypass capacitor can go from 330 nF to 1 uF.
6.2 Inductor selection
The inductance value fixes the current ripple flowing through the output capacitor. So the
minimum inductance value to have the expected current ripple must be selected. The rule to
fix the current ripple value is to have a ripple at 20%-40% of the output current.
In continuous current mode (CCM), the inductance value can be calculated by the following
equation:
Equation 19
where TON is the conduction time of the high-side switch and TOFF is the conduction time of
the low-side switch (in CCM, FSW=1/(TON + TOFF)). The maximum current ripple, given the
Vout, is obtained at maximum TOFF, that is at minimum duty cycle (see previous section to
calculate minimum duty). So, fixing ΔIL=20% to 30% of the maximum output current, the
minimum inductance value can be calculated as:
Equation 20
where FSWMIN is the minimum switching frequency, according to Table 4.
The peak current through the inductor is given by:
Equation 21
Table 6. Input MLCC capacitors
Manufacturer Series Cap value (μF) Rated voltage (V)
Murata GRM31 10 25
GRM55 10 25
TDK C3225 10 25
ΔIL
VIN VOUT
L
------------------------------TON
VOUT
L
-------------- TOFF
==
LMIN
VOUT
ΔIMAX
---------------- 1D
MIN
FSWMIN
-----------------------
=
ILPK,IO
ΔIL
2
--------+=
é]
DocID023654 Rev 2 17/27
ST1S41 Application information
So, if the inductor value decreases, the peak current (that must be lower than the current
limit of the device) increases. The higher the inductor value, the higher the average output
current that can be delivered, without reaching the current limit.
In Table 7 some inductor part numbers are listed.
6.3 Output capacitor selection
The current in the output capacitor has a triangular waveform which generates a voltage
ripple across it. This ripple is due to the capacitive component (charge or discharge of the
output capacitor) and the resistive component (due to the voltage drop across its ESR). So
the output capacitor must be selected in order to have a voltage ripple compliant with the
application requirements.
The amount of the voltage ripple can be calculated starting from the current ripple obtained
by the inductor selection.
Equation 22
For the ceramic capacitor (MLCC) the capacitive component of the ripple dominates the
resistive one. While for the electrolythic capacitor the opposite is true.
Since the compensation network is internal, the output capacitor should be selected in order
to have a proper phase margin and then a stable control loop.
The equations of Section 5.2 help to check loop stability given the application conditions,
the value of the inductor, and of the output capacitor.
In Table 8 some capacitor series are listed.
Table 7. Inductors
Manufacturer Series Inductor value (μH) Saturation current (A)
Coilcraft
XAL5030/6030 2.2 to 4.7 6.7 to 15.5
MSS1048 2.2 to 6.8 4.14 to 6.62
MSS1260 10 5.5
Wurth
WE-HC/HCA 3.3 to 4.7 7 to 11
WE-TPC typ XLH 3.6 to 6.2 4.5 to 6.4
WE-PD type L 10 5.6
TDK RLF7030T 2.2 to 4.7 4 to 6
Table 8. Output capacitors
Manufacturer Series Cap value (μF) Rated voltage (V) ESR (mΩ)
MURATA GRM32 22 to 100 6.3 to 25 < 5
GRM31 10 to 47 6.3 to 25 < 5
ΔVOUT ESR ΔIMAX
ΔIMAX
8C
OUT fSW
⋅⋅
------------------------------------------+=
Application information ST1S41
18/27 DocID023654 Rev 2
6.4 Thermal dissipation
The thermal design is important to prevent the thermal shutdown of the device if junction
temperature goes above 150 °C. The three different sources of losses within the device are:
a) conduction losses due to the on-resistance of the high-side switch (RHS) and low-
side switch (RLS); these are equal to:
Equation 23
where D is the duty cycle of the application. Note that the duty cycle is theoretically given by
the ratio between VOUT and VIN, but actually it is slightly higher to compensate the losses of
the regulator.
b) switching losses due to high-side power MOSFET turn-on and off; these can be
calculated as:
Equation 24
where TRISE and TFALL are the overlap times of the voltage across the high-side power
switch (VDS) and the current flowing into it during turn-on and turn-off phases, as shown in
Figure 7. TSW is the equivalent switching time. For this device the typical value for the
equivalent switching time is 20 ns.
c) Quiescent current losses, calculated as:
Equation 25
where IQ is the quiescent current (IQ=2.5 mA maximum).
The junction temperature TJ can be calculated as:
PANASONIC ECJ 10 to 22 6.3 < 5
EEFCD 10 to 68 6.3 15 to 55
SANYO TPA/B/C 100 to 470 4 to 16 40 to 80
TDK C3225 22 to 100 6.3 < 5
Table 8. Output capacitors (continued)
Manufacturer Series Cap value (μF) Rated voltage (V) ESR (mΩ)
PCOND RHS IOUT2DR
LS IOUT21D()⋅⋅+⋅⋅=
PSW VIN IOUT
TRISE TFALL
+()
2
-------------------------------------------Fsw⋅⋅ VIN IOUT TSW FSW
⋅⋅==
PQVIN IQ
=
DocID023654 Rev 2 19/27
ST1S41 Application information
Equation 26
where TA is the ambient temperature and PTOT is the sum of the power losses just seen.
RthJA is the equivalent thermal resistance junction-to-ambient of the device; it can be
calculated as the parallel of many paths of heat conduction from the junction to the ambient.
For this device the path through the exposed pad is the one conducting the largest amount
of heat. The RthJA measured on the demonstration board described in the following
paragraph is about 40 °C/W for the VFQFPN and HSOP packages.
Figure 7. Switching losses
6.5 Layout considerations
The PC board layout of switching DC-DC regulator is very important to minimize the noise
injected in high impedance nodes, to reduce interference generated by the high switching
current loops and to optimize the reliability of the device.
In order to avoid EMC problems, the high switching current loops must be as short as
possible. In the buck converter there are two high switching current loops: during the on-
time, the pulsed current flows through the input capacitor, the high-side power switch, the
inductor and the output capacitor; during the off-time, through the low-side power switch, the
inductor and the output capacitor.
TJTARthJA PTOT
+=
Application information ST1S41
20/27 DocID023654 Rev 2
The input capacitor connected to VINSW must be placed as close as possible to the device,
to avoid spikes on VINSW due to the stray inductance and the pulsed input current.
In order to prevent dynamic unbalance between VINSW and VINA, the trace connecting the
VINA pin to the input must be derived from VINSW.
The feedback pin (FB) connection to the external resistor divider is a high impedance node,
so the interference can be minimized routing the feedback node with a very short trace and
as far as possible from the high current paths.
A single point connection from signal ground to power ground is suggested.
Thanks to the exposed pad of the device, the ground plane helps to reduce the thermal
resistance junction-to-ambient; so a large ground plane, soldered to the exposed pad,
enhances the thermal performance of the converter allowing high power conversion.
Figure 8. Suggested PCB layout
Input cap as close as possible
t
o VINSW pin
Star center for common ground
Short FB trace VINA derived from Cin
to avoid dynamic voltage drop
between VINA and VINSW
Short high switching
current loop
Via to connect the thermal pad
To bottom or inner ground plane
AM15065v1
OUY OUT
DocID023654 Rev 2 21/27
ST1S41 Typical characteristics
7 Typical characteristics
Figure 9. Efficiency vs. IOUT@Vin=5 V Figure 10. Efficiency vs. IOUT@Vin=12 V
Figure 11. Start at full load 4 A Figure 12. Efficiency vs. IOUT@ different Vo
values
40
50
60
70
80
90
100
0.00 0.50 1.00 1.50 2.00 2.50 3.00 3.50 4.00 4.50
Efficiency [%]
Iout [A]
Vin=5V
Vo=3.3V
Vo=1.8V
Vo=1.2V
AM15066v1
40
50
60
70
80
90
100
0.00 0.50 1.00 1.50 2.00 2.50 3.00 3.50 4.00 4.50
Efficiency [%]
Iout [A]
Vin=12V
Vo=5V
Vo=3.3V
AM15067v1
Vout
IL
4A/div
200mV/div
200us/div
AM15068v1
40
45
50
55
60
65
70
75
80
85
90
0.00 0.50 1.00 1.50 2.00 2.50 3.00 3.50 4.00 4.50
Efficiency [%]
Iout [A]
Vin=12V
Vo=1.8V
Vo=1.2V
AM15069v1
Typical characteristics ST1S41
22/27 DocID023654 Rev 2
Figure 13. Startup with output shorted
Vout
IL
2A / d i v
5V/div
200us/div
SW
10us/div
AM14070v1
é] 75695533
DocID023654 Rev 2 23/27
ST1S41 Package mechanical data
8 Package mechanical data
In order to meet environmental requirements, ST offers these devices in different grades of
ECOPACK® packages, depending on their level of environmental compliance. ECOPACK®
specifications, grade definitions and product status are available at: www.st.com.
ECOPACK® is an ST trademark.
Figure 14. VFQFPN8 (4x4x1.08 mm) package dimensions
Table 9. VFQFPN8 (4x4x1.08 mm) mechanical data
Dim. mm
Min. Typ. Max.
A 0.80 0.90 1.00
A1 0.02 0.05
A3 0.20
b 0.23 0.30 0.38
D 3.90 4.00 4.10
D2 2.82 3.00 3.23
E 3.90 4.00 4.10
E2 2.05 2.20 2.30
e0.80
L 0.40 0.50 0.60
m4 u n: “In W AMI 1755“ l7
Package mechanical data ST1S41
24/27 DocID023654 Rev 2
Figure 15. HSOP8 package dimensions
Table 10. HSOP8 mechanical data
Dim. mm
Min. Typ. Max.
A1.70
A1 0.00 0.15
A2 1.25
b 0.31 0.51
c 0.17 0.25
D 4.80 4.90 5.00
E 5.80 6.00 6.20
E1 3.80 3.90 4.00
e1.27
h 0.25 0.50
L 0.40 1.27
k0 8
ccc 0.10
E]
DocID023654 Rev 2 25/27
ST1S41 Ordering information
9 Ordering information
Table 11. Ordering information
Order code Package
ST1S41PUR VFQFPN 4x4 8L
ST1S41PHR HSOP8
Revision history ST1S41
26/27 DocID023654 Rev 2
10 Revision history
Table 12. Document revision history
Date Revision Changes
14-Sep-2012 1Initial release.
24-Apr-2013 2 Updated Table 4: Electrical characteristics and Table 11: Ordering
information.
é]
DocID023654 Rev 2 27/27
ST1S41
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